System and methods to compensate for doppler effects in multi-user (MU) multiple antenna systems (MAS)

ABSTRACT

A system and methods are described which compensate for the adverse effect of Doppler on the performance of DIDO systems. One embodiment of such a system employs different selection algorithms to adaptively adjust the active BTSs to different UEs based by tracking the changing channel conditions. Another embodiment utilizes channel prediction to estimate the future CSI or DIDO precoding weights, thereby eliminating errors due to outdated CSI.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.13/464,648, filed May 4, 2012, entitled, “System And Methods ToCompensate For Doppler Effects In Mulit-User (MU) Antenna Systems (MAS)”which is a continuation-in-part of the following co-pending patentapplications:

U.S. application Ser. No. 12/917,257, filed Nov. 1, 2010, entitled“Systems And Methods To Coordinate Transmissions In Distributed WirelessSystems Via User Clustering”

U.S. application Ser. No. 12/802,988, filed Jun. 16, 2010, entitled“Interference Management, Handoff, Power Control And Link Adaptation InDistributed-Input Distributed-Output (DIDO) Communication Systems”

U.S. application Ser. No. 12/802,976, filed Jun. 16, 2010, entitled“System And Method For Adjusting DIDO Interference Cancellation Based OnSignal Strength Measurements”

U.S. application Ser. No. 12/802,974, filed Jun. 16, 2010, entitled“System And Method For Managing Inter-Cluster Handoff Of Clients WhichTraverse Multiple DIDO Clusters”

U.S. application Ser. No. 12/802,989, filed Jun. 16, 2010, entitled“System And Method For Managing Handoff Of A Client Between DifferentDistributed-Input-Distributed-Output (DIDO) Networks Based On DetectedVelocity Of The Client”

U.S. application Ser. No. 12/802,958, filed Jun. 16, 2010, entitled“System And Method For Power Control And Antenna Grouping In ADistributed-Input-Distributed-Output (DIDO) Network”

U.S. application Ser. No. 12/802,975, filed Jun. 16, 2010, entitled“System And Method For Link adaptation In DIDO Multicarrier Systems”

U.S. application Ser. No. 12/802,938, filed Jun. 16, 2010, entitled“System And Method For DIDO Precoding Interpolation In MulticarrierSystems”

U.S. application Ser. No. 12/630,627, filed Dec. 3, 2009, entitled“System and Method For Distributed Antenna Wireless Communications”

U.S. application Ser. No. 12/143,503, filed Jun. 20, 2008 entitled“System and Method For Distributed Input-Distributed Output WirelessCommunications”;

U.S. application Ser. No. 11/894,394, filed Aug. 20, 2007 entitled,“System and Method for Distributed Input Distributed Output WirelessCommunications”;

U.S. application Ser. No. 11/894,362, filed Aug. 20, 2007 entitled,“System and method for Distributed Input-Distributed WirelessCommunications”;

U.S. application Ser. No. 11/894,540, filed Aug. 20, 2007 entitled“System and Method For Distributed Input-Distributed Output WirelessCommunications”

U.S. application Ser. No. 11/256,478, filed Oct. 21, 2005 entitled“System and Method For Spatial-Multiplexed Tropospheric ScatterCommunications”;

U.S. application Ser. No. 10/817,731, filed Apr. 2, 2004 entitled“System and Method For Enhancing Near Vertical Incidence Skywave(“NVIS”) Communication Using Space-Time Coding.

BACKGROUND

Prior art multi-user wireless systems may include only a single basestation or several base stations.

A single WiFi base station (e.g., utilizing 2.4 GHz 802.11b, g or nprotocols) attached to a broadband wired Internet connection in an areawhere there are no other WiFi access points (e.g. a WiFi access pointattached to DSL within a rural home) is an example of a relativelysimple multi-user wireless system that is a single base station that isshared by one or more users that are within its transmission range. If auser is in the same room as the wireless access point, the user willtypically experience a high-speed link with few transmission disruptions(e.g. there may be packet loss from 2.4 GHz interferers, like microwaveovens, but not from spectrum sharing with other WiFi devices), If a useris a medium distance away or with a few obstructions in the path betweenthe user and WiFi access point, the user will likely experience amedium-speed link. If a user is approaching the edge of the range of theWiFi access point, the user will likely experience a low-speed link, andmay be subject to periodic drop-outs if changes to the channel result inthe signal SNR dropping below usable levels. And, finally, if the useris beyond the range of the WiFi base station, the user will have no linkat all.

When multiple users access the WiFi base station simultaneously, thenthe available data throughput is shared among them. Different users willtypically place different throughput demands on a WiFi base station at agiven time, but at times when the aggregate throughput demands exceedthe available throughput from the WiFi base station to the users, thensome or all users will receive less data throughput than they areseeking. In an extreme situation where a WiFi access point is sharedamong a very large number of users, throughput to each user can slowdown to a crawl, and worse, data throughput to each user may arrive inshort bursts separated by long periods of no data throughput at all,during which time other users are served. This “choppy” data deliverymay impair certain applications, like media streaming.

Adding additional WiFi base stations in situations with a large numberof users will only help up to a point. Within the 2.4 GHz ISM band inthe U.S., there are 3 non-interfering channels that can be used forWiFi, and if 3 WiFi base stations in the same coverage area areconfigured to each use a different non-interfering channel, then theaggregate throughput of the coverage area among multiple users will beincreased up to a factor of 3. But, beyond that, adding more WiFi basestations in the same coverage area will not increase aggregatethroughput, since they will start sharing the same available spectrumamong them, effectually utilizing time-division multiplexed access(TDMA) by “taking turns” using the spectrum. This situation is oftenseen in coverage areas with high population density, such as withinmulti-dwelling units. For example, a user in a large apartment buildingwith a WiFi adapter may well experience very poor throughput due todozens of other interfering WiFi networks (e.g. in other apartments)serving other users that are in the same coverage area, even if theuser's access point is in the same room as the client device accessingthe base station. Although the link quality is likely good in thatsituation, the user would be receiving interference from neighbor WiFiadapters operating in the same frequency band, reducing the effectivethroughput to the user.

Current multiuser wireless systems, including both unlicensed spectrum,such as WiFi, and licensed spectrum, suffer from several limitations.These include coverage area, downlink (DL) data rate and uplink (UL)data rate. Key goals of next generation wireless systems, such as WiMAXand LTE, are to improve coverage area and DL and UL data rate viamultiple-input multiple-output (MIMO) technology. MIMO employs multipleantennas at transmit and receive sides of wireless links to improve linkquality (resulting in wider coverage) or data rate (by creating multiplenon-interfering spatial channels to every user). If enough data rate isavailable for every user (note, the terms “user” and “client” are usedherein interchangeably), however, it may be desirable to exploit channelspatial diversity to create non-interfering channels to multiple users(rather than single user), according to multiuser MIMO (MU-MIMO)techniques. See, e.g., the following references:

-   G. Caire and S. Shamai, “On the achievable throughput of a    multiantenna Gaussian broadcast channel,” IEEE Trans. Info. Th.,    vol. 49, pp. 1691-1706, July 2003.-   P. Viswanath and D. Tse, “Sum capacity of the vector Gaussian    broadcast channel and uplink-downlink duality,” IEEE Trans. Info.    Th., vol. 49, pp. 1912-1921, August 2003.-   S. Vishwanath, N. Jindal, and A. Goldsmith, “Duality, achievable    rates, and sum-rate capacity of Gaussian MIMO broadcast channels,”    IEEE Trans. Info. Th., vol. 49, pp. 2658-2668, October 2003.-   W. Yu and J. Cioffi, “Sum capacity of Gaussian vector broadcast    channels,” IEEE Trans. Info. Th., vol. 50, pp. 1875-1892, September    2004.-   M. Costa, “Writing on dirty paper,” IEEE Transactions on Information    Theory, vol. 29, pp. 439-441, May 1983.-   M. Bengtsson, “A pragmatic approach to multi-user spatial    multiplexing,” Proc. of Sensor Array and Multichannel Sign. Proc.    Workshop, pp. 130-134, August 2002.-   K.-K. Wong, R. D. Murch, and K. B. Letaief, “Performance enhancement    of multiuser MIMO wireless communication systems,” IEEE Trans.    Comm., vol. 50, pp. 1960-1970, December 2002.-   M. Sharif and B. Hassibi, “On the capacity of MIMO broadcast channel    with partial side information,” IEEE Trans. Info. Th., vol. 51, pp.    506-522, February 2005.

For example, in MIMO 4×4 systems (i.e., four transmit and four receiveantennas), 10 MHz bandwidth, 16-QAM modulation and forward errorcorrection (FEC) coding with rate 3/4 (yielding spectral efficiency of 3bps/Hz), the ideal peak data rate achievable at the physical layer forevery user is 4×30 Mbps=120 Mbps, which is much higher than required todeliver high definition video content (which may only require ˜10 Mbps).In MU-MIMO systems with four transmit antennas, four users and singleantenna per user, in ideal scenarios (i.e., independent identicallydistributed, i.i.d., channels) downlink data rate may be shared acrossthe four users and channel spatial diversity may be exploited to createfour parallel 30 Mbps data links to the users. Different MU-MIMO schemeshave been proposed as part of the LTE standard as described, forexample, in 3GPP, “Multiple Input Multiple Output in UTRA”, 3GPP TR25.876 V7.0.0, March 2007; 3GPP, “Base Physical channels andmodulation”, TS 36.211, V8.7.0, May 2009; and 3GPP, “Multiplexing andchannel coding”, TS 36.212, V8.7.0, May 2009. However, these schemes canprovide only up to 2× improvement in DL data rate with four transmitantennas. Practical implementations of MU-MIMO techniques in standardand proprietary cellular systems by companies like ArrayComm (see, e.g.,ArrayComm, “Field-proven results”,http://www.arraycomm.com/serve.php?page=proof) have yielded up to a ˜3×increase (with four transmit antennas) in DL data rate via spacedivision multiple access (SDMA). A key limitation of MU-MIMO schemes incellular networks is lack of spatial diversity at the transmit side.Spatial diversity is a function of antenna spacing and multipath angularspread in the wireless links. In cellular systems employing MU-MIMOtechniques, transmit antennas at a base station are typically clusteredtogether and placed only one or two wavelengths apart due to limitedreal estate on antenna support structures (referred to herein as“towers,” whether physically tall or not) and due to limitations onwhere towers may be located. Moreover, multipath angular spread is lowsince cell towers are typically placed high up (10 meters or more) aboveobstacles to yield wider coverage.

Other practical issues with cellular system deployment include excessivecost and limited availability of locations for cellular antennalocations (e.g. due to municipal restrictions on antenna placement, costof real-estate, physical obstructions, etc.) and the cost and/oravailability of network connectivity to the transmitters (referred toherein as “backhaul”). Further, cellular systems often have difficultyreaching clients located deeply in buildings due to losses from walls,ceilings, floors, furniture and other impediments.

Indeed, the entire concept of a cellular structure for wide-area networkwireless presupposes a rather rigid placement of cellular towers, analternation of frequencies between adjacent cells, and frequentlysectorization, so as to avoid interference among transmitters (eitherbase stations or users) that are using the same frequency. As a result,a given sector of a given cell ends up being a shared block of DL and ULspectrum among all of the users in the cell sector, which is then sharedamong these users primarily in only the time domain. For example,cellular systems based on Time Division Multiple Access (TDMA) and CodeDivision Multiple Access (CDMA) both share spectrum among users in thetime domain. By overlaying such cellular systems with sectorization,perhaps a 2-3× spatial domain benefit can be achieved. And, then byoverlaying such cellular systems with a MU-MIMO system, such as thosedescribed previously, perhaps another 2-3× space-time domain benefit canbe achieved. But, given that the cells and sectors of the cellularsystem are typically in fixed locations, often dictated by where towerscan be placed, even such limited benefits are difficult to exploit ifuser density (or data rate demands) at a given time does not match upwell with tower/sector placement. A cellular smart phone user oftenexperiences the consequence of this today where the user may be talkingon the phone or downloading a web page without any trouble at all, andthen after driving (or even walking) to a new location will suddenly seethe voice quality drop or the web page slow to a crawl, or even lose theconnection entirely. But, on a different day, the user may have theexact opposite occur in each location. What the user is probablyexperiencing, assuming the environmental conditions are the same, is thefact that user density (or data rate demands) is highly variable, butthe available total spectrum (and thereby total data rate, using priorart techniques) to be shared among users at a given location is largelyfixed.

Further, prior art cellular systems rely upon using differentfrequencies in different adjacent cells, typically 3 differentfrequencies. For a given amount of spectrum, this reduces the availabledata rate by 3×.

So, in summary, prior art cellular systems may lose perhaps 3× inspectrum utilization due to cellularization, and may improve spectrumutilization by perhaps 3× through sectorization and perhaps 3× morethrough MU-MIMO techniques, resulting in a net 3*3/3=3× potentialspectrum utilization. Then, that bandwidth is typically divided up amongusers in the time domain, based upon what sector of what cell the usersfall into at a given time. There are even further inefficiencies thatresult due to the fact that a given user's data rate demands aretypically independent of the user's location, but the available datarate varies depending on the link quality between the user and the basestation. For example, a user further from a cellular base station willtypically have less available data rate than a user closer to a basestation. Since the data rate is typically shared among all of the usersin a given cellular sector, the result of this is that all users areimpacted by high data rate demands from distant users with poor linkquality (e.g. on the edge of a cell) since such users will still demandthe same amount of data rate, yet they will be consuming more of theshared spectrum to get it.

Other proposed spectrum sharing systems, such as that used by WiFi(e.g., 802.11b, g, and n) and those proposed by the White SpacesCoalition, share spectrum very inefficiently since simultaneoustransmissions by base stations within range of a user result ininterference, and as such, the systems utilize collision avoidance andsharing protocols. These spectrum sharing protocols are within the timedomain, and so, when there are a large number of interfering basestations and users, no matter how efficient each base station itself isin spectrum utilization, collectively the base stations are limited totime domain sharing of the spectrum among each other. Other prior artspectrum sharing systems similarly rely upon similar methods to mitigateinterference among base stations (be they cellular base stations withantennas on towers or small scale base stations, such as WiFi AccessPoints (APs)). These methods include limiting transmission power fromthe base station so as to limit the range of interference, beamforming(via synthetic or physical means) to narrow the area of interference,time-domain multiplexing of spectrum and/or MU-MIMO techniques withmultiple clustered antennas on the user device, the base station orboth. And, in the case of advanced cellular networks in place or plannedtoday, frequently many of these techniques are used at once.

But, what is apparent by the fact that even advanced cellular systemscan achieve only about a 3× increase in spectrum utilization compared toa single user utilizing the spectrum is that all of these techniqueshave done little to increase the aggregate data rate among shared usersfor a given area of coverage. In particular, as a given coverage areascales in terms of users, it becomes increasingly difficult to scale theavailable data rate within a given amount of spectrum to keep pace withthe growth of users. For example, with cellular systems, to increase theaggregate data rate within a given area, typically the cells aresubdivided into smaller cells (often called nano-cells or femto-cells).Such small cells can become extremely expensive given the limitations onwhere towers can be placed, and the requirement that towers must beplaced in a fairly structured pattern so as to provide coverage with aminimum of “dead zones”, yet avoid interference between nearby cellsusing the same frequencies. Essentially, the coverage area must bemapped out, the available locations for placing towers or base stationsmust be identified, and then given these constraints, the designers ofthe cellular system must make do with the best they can. And, of course,if user data rate demands grow over time, then the designers of thecellular system must yet again remap the coverage area, try to findlocations for towers or base stations, and once again work within theconstraints of the circumstances. And, very often, there simply is nogood solution, resulting in dead zones or inadequate aggregate data ratecapacity in a coverage area. In other words, the rigid physicalplacement requirements of a cellular system to avoid interference amongtowers or base stations utilizing the same frequency results insignificant difficulties and constraints in cellular system design, andoften is unable to meet user data rate and coverage requirements.

So-called prior art “cooperative” and “cognitive” radio systems seek toincrease the spectral utilization in a given area by using intelligentalgorithms within radios such that they can minimize interference amongeach other and/or such that they can potentially “listen” for otherspectrum use so as to wait until the channel is clear. Such systems areproposed for use particularly in unlicensed spectrum in an effort toincrease the spectrum utilization of such spectrum.

A mobile ad hoc network (MANET) (seehttp://en.wikipedia.org/wiki/Mobile_ad_hoc_network) is an example of acooperative self-configuring network intended to provide peer-to-peercommunications, and could be used to establish communication amongradios without cellular infrastructure, and with sufficiently low-powercommunications, can potentially mitigate interference among simultaneoustransmissions that are out of range of each other. A vast number ofrouting protocols have been proposed and implemented for MANET systems(see http://en.wikipedia.org/wiki/List_of_ad-hoc_routing_protocols for alist of dozens of routing protocols in a wide range of classes), but acommon theme among them is they are all techniques for routing (e.g.repeating) transmissions in such a way to minimize transmitterinterference within the available spectrum, towards the goal ofparticular efficiency or reliability paradigms.

All of the prior art multi-user wireless systems seek to improvespectrum utilization within a given coverage area by utilizingtechniques to allow for simultaneous spectrum utilization among basestations and multiple users. Notably, in all of these cases, thetechniques utilized for simultaneous spectrum utilization among basestations and multiple users achieve the simultaneous spectrum use bymultiple users by mitigating interference among the waveforms to themultiple users. For example, in the case of 3 base stations each using adifferent frequency to transmit to one of 3 users, there interference ismitigated because the 3 transmissions are at 3 different frequencies. Inthe case of sectorization from a base station to 3 different users, each180 degrees apart relative to the base station, interference ismitigated because the beamforming prevents the 3 transmissions fromoverlapping at any user.

When such techniques are augmented with MU-MIMO, and, for example, eachbase station has 4 antennas, then this has the potential to increasedownlink throughput by a factor of 4, by creating four non-interferingspatial channels to the users in given coverage area. But it is stillthe case that some technique must be utilized to mitigate theinterference among multiple simultaneous transmissions to multiple usersin different coverage areas.

And, as previously discussed, such prior art techniques (e.g.cellularization, sectorization) not only typically suffer fromincreasing the cost of the multi-user wireless system and/or theflexibility of deployment, but they typically run into physical orpractical limitations of aggregate throughput in a given coverage area.For example, in a cellular system, there may not be enough availablelocations to install more base stations to create smaller cells. And, inan MU-MIMO system, given the clustered antenna spacing at each basestation location, the limited spatial diversity results inasymptotically diminishing returns in throughput as more antennas areadded to the base station.

And further, in the case of multi-user wireless systems where the userlocation and density is unpredictable, it results in unpredictable (withfrequently abrupt changes) in throughput, which is inconvenient to theuser and renders some applications (e.g. the delivery of servicesrequiring predictable throughput) impractical or of low quality. Thus,prior art multi-user wireless systems still leave much to be desired interms of their ability to provide predictable and/or high-qualityservices to users.

Despite the extraordinary sophistication and complexity that has beendeveloped for prior art multi-user wireless systems over time, thereexist common themes: transmissions are distributed among different basestations (or ad hoc transceivers) and are structured and/or controlledso as to avoid the RF waveform transmissions from the different basestations and/or different ad hoc transceivers from interfering with eachother at the receiver of a given user.

Or, to put it another way, it is taken as a given that if a user happensto receive transmissions from more than one base station or ad hoctransceiver at the same time, the interference from the multiplesimultaneous transmissions will result in a reduction of the SNR and/orbandwidth of the signal to the user which, if severe enough, will resultin loss of all or some of the potential data (or analog information)that would otherwise have been received by the user.

Thus, in a multiuser wireless system, it is necessary to utilize one ormore spectrum sharing approaches or another to avoid or mitigate suchinterference to users from multiple base stations or ad hoc transceiverstransmitting at the same frequency at the same time. There are a vastnumber of prior art approaches to avoiding such interference, includingcontrolling base stations' physical locations (e.g. cellularization),limiting power output of base stations and/or ad hoc transceivers (e.g.limiting transmit range), beamforming/sectorization, and time domainmultiplexing. In short, all of these spectrum sharing systems seek toaddress the limitation of multiuser wireless systems that when multiplebase stations and/or ad hoc transceivers transmitting simultaneously atthe same frequency are received by the same user, the resultinginterference reduces or destroys the data throughput to the affecteduser. If a large percentage, or all, of the users in the multi-userwireless system are subject to interference from multiple base stationsand/or ad hoc transceivers (e.g. in the event of the malfunction of acomponent of a multi-user wireless system), then it can result in asituation where the aggregate throughput of the multi-user wirelesssystem is dramatically reduced, or even rendered non-functional.

Prior art multi-user wireless systems add complexity and introducelimitations to wireless networks and frequently result in a situationwhere a given user's experience (e.g. available bandwidth, latency,predictability, reliability) is impacted by the utilization of thespectrum by other users in the area. Given the increasing demands foraggregate bandwidth within wireless spectrum shared by multiple users,and the increasing growth of applications that can rely upon multi-userwireless network reliability, predictability and low latency for a givenuser, it is apparent that prior art multi-user wireless technologysuffers from many limitations. Indeed, with the limited availability ofspectrum suitable for particular types of wireless communications (e.g.at wavelengths that are efficient in penetrating building walls), it maybe the case that prior art wireless techniques will be insufficient tomeet the increasing demands for bandwidth that is reliable, predictableand low-latency.

Prior art related to the current invention describes beamforming systemsand methods for null-steering in multiuser scenarios. Beamforming wasoriginally conceived to maximize received signal-to-noise ratio (SNR) bydynamically adjusting phase and/or amplitude of the signals (i.e.,beamforming weights) fed to the antennas of the array, thereby focusingenergy toward the user's direction. In multiuser scanarios, beamformingcan be used to suppress interfering sources and maximizesignal-to-interference-plus-noise ratio (SINR). For example, whenbeamforming is used at the receiver of a wireless link, the weights arecomputed to create nulls in the direction of the interfering sources.When beamforming is used at the transmitter in multiuser downlinkscenarios, the weights are calculated to pre-cancel inter-userinterference and maximize the SINR to every user. Alternative techniquesfor multiuser systems, such as BD precoding, compute the precodingweights to maximize throughput in the downlink broadcast channel. Theco-pending applications, which are incorporated herein by reference,describe the foregoing techniques (see co-pending applications forspecific citations).

BRIEF DESCRIPTION OF THE DRAWINGS

A better understanding of the present invention can be obtained from thefollowing detailed description in conjunction with the drawings, inwhich:

FIG. 1 illustrates a main DIDO cluster surrounded by neighboring DIDOclusters in one embodiment of the invention.

FIG. 2 illustrates frequency division multiple access (FDMA) techniquesemployed in one embodiment of the invention.

FIG. 3 illustrates time division multiple access (TDMA) techniquesemployed in one embodiment of the invention.

FIG. 4 illustrates different types of interfering zones addressed in oneembodiment of the invention.

FIG. 5 illustrates a framework employed in one embodiment of theinvention.

FIG. 6 illustrates a graph showing SER as a function of the SNR,assuming SIR=10 dB for the target client in the interfering zone.

FIG. 7 illustrates a graph showing SER derived from two IDCI-precodingtechniques.

FIG. 8 illustrates an exemplary scenario in which a target client movesfrom a main DIDO cluster to an interfering cluster.

FIG. 9 illustrates the signal-to-interference-plus-noise ratio (SINR) asa function of distance (D).

FIG. 10 illustrates the symbol error rate (SER) performance of the threescenarios for 4-QAM modulation in flat-fading narrowband channels.

FIG. 11 illustrates a method for IDCI precoding according to oneembodiment of the invention.

FIG. 12 illustrates the SINR variation in one embodiment as a functionof the client's distance from the center of main DIDO clusters.

FIG. 13 illustrates one embodiment in which the SER is derived for 4-QAMmodulation.

FIG. 14 illustrates one embodiment of the invention in which a finitestate machine implements a handoff algorithm.

FIG. 15 illustrates depicts one embodiment of a handoff strategy in thepresence of shadowing.

FIG. 16 illustrates a hysteresis loop mechanism when switching betweenany two states in FIG. 93.

FIG. 17 illustrates one embodiment of a DIDO system with power control.

FIG. 18 illustrates the SER versus SNR assuming four DIDO transmitantennas and four clients in different scenarios.

FIG. 19 illustrates MPE power density as a function of distance from thesource of RF radiation for different values of transmit power accordingto one embodiment of the invention.

FIGS. 20a-b illustrate different distributions of low-power andhigh-power DIDO distributed antennas.

FIGS. 21a-b illustrate two power distributions corresponding to theconfigurations in FIGS. 20a and 20b , respectively.

FIG. 22a-b illustrate the rate distribution for the two scenarios shownin FIGS. 99a and 99b , respectively.

FIG. 23 illustrates one embodiment of a DIDO system with power control.

FIG. 24 illustrates one embodiment of a method which iterates across allantenna groups according to Round-Robin scheduling policy fortransmitting data.

FIG. 25 illustrates a comparison of the uncoded SER performance of powercontrol with antenna grouping against conventional eigenmode selectionin U.S. Pat. No. 7,636,381.

FIGS. 26a-c illustrate three scenarios in which BD precoding dynamicallyadjusts the precoding weights to account for different power levels overthe wireless links between DIDO antennas and clients.

FIG. 27 illustrates the amplitude of low frequency selective channels(assuming β=1) over delay domain or instantaneous PDP (upper plot) andfrequency domain (lower plot) for DIDO 2×2 systems

FIG. 28 illustrates one embodiment of a channel matrix frequencyresponse for DIDO 2×2, with a single antenna per client.

FIG. 29 illustrates one embodiment of a channel matrix frequencyresponse for DIDO 2×2, with a single antenna per client for channelscharacterized by high frequency selectivity (e.g., with β=0.1).

FIG. 30 illustrates exemplary SER for different QAM schemes (i.e.,4-QAM, 16-QAM, 64-QAM).

FIG. 31 illustrates one embodiment of a method for implementing linkadaptation (LA) techniques.

FIG. 32 illustrates SER performance of one embodiment of the linkadaptation (LA) techniques.

FIG. 33 illustrates the entries of the matrix in equation (28) as afunction of the OFDM tone index for DIDO 2×2 systems with N_(FFT)=64 andL₀=8.

FIG. 34 illustrates the SER versus SNR for L₀=8, M=N_(t)=2 transmitantennas and a variable number of P.

FIG. 35 illustrates the SER performance of one embodiment of aninterpolation method for different DIDO orders and L₀=16.

FIG. 36 illustrates one embodiment of a system which employssuper-clusters, DIDO-clusters and user-clusters.

FIG. 37 illustrates a system with user clusters according to oneembodiment of the invention.

FIGS. 38a-b illustrate link quality metric thresholds employed in oneembodiment of the invention.

FIGS. 39-41 illustrate examples of link-quality matrices forestablishing user clusters.

FIG. 42 illustrates an embodiment in which a client moves acrossdifferent different DIDO clusters.

FIGS. 43-46 illustrate relationships between the resolution of sphericalarrays and their area A in one embodiment of the invention.

FIG. 47 illustrates the degrees of freedom of MIMO systems in practicalindoor and outdoor propagation scenarios.

FIG. 48 illustrates the degrees of freedom in DIDO systems as a functionof the array diameter.

FIG. 49 illustrates one embodiment which includes multiple centralizedprocessors (CP) and distributed nodes (DN) that communicate via wirelineor wireless connections.

FIG. 50 illustrates one embodiment in which CPs exchange controlinformation with the unlicensed DNs and reconfigure them to shut downthe frequency bands for licensed use.

FIG. 51 illustrates one embodiment in which an entire spectrum isallocated to the new service and control information is used by the CPsto shut down all unlicensed DNs to avoid interference with the licensedDNs.

FIG. 52 illustrates one embodiment of a cloud wireless system includingmultiple CPs, distributed nodes and a network interconnecting the CPs tothe DNs.

FIGS. 53-59 illustrate embodiments of a multiuser (MU) multiple antennasystem (MAS) that adaptively reconfigures parameters to compensate forDoppler effects due to user mobility or changes in the propagationenvironment.

FIG. 60 illustrates a plurality of BTSs, some of which have good SNR andsome of which have low Doppler with respect to a UE.

FIG. 61 illustrates one embodiment of a matrix containing values of SNRand Doppler recorded by a CP for a plurality of BTS-UE links.

FIG. 62 illustrates the channel gain (or CSI) at different times inaccordance with one embodiment of the invention.

DETAILED DESCRIPTION

One solution to overcome many of the above prior art limitations is anembodiment of Distributed-Input Distributed-Output (DIDO) technology.DIDO technology is described in the following patents and patentapplications, all of which are assigned the assignee of the presentpatent and are incorporated by reference. The present application is acontinuation in part (CIP) to these patent applications. These patentsand applications are sometimes referred to collectively herein as the“related patents and applications”:

U.S. application Ser. No. 13/232,996, filed Sep. 14, 2011, entitled“Systems And Methods To Exploit Areas of Coherence in Wirless Systems”

U.S. application Ser. No. 13/233,006, filed Sep. 14, 2011, entitled“Systems and Methods for Planned Evoluation and Obsolescence ofMultiuser Spectrum.”

U.S. application Ser. No. 12/917,257, filed Nov. 1, 2010, entitled“Systems And Methods To Coordinate Transmissions In Distributed WirelessSystems Via User Clustering”

U.S. application Ser. No. 12/802,988, filed Jun. 16, 2010, entitled“Interference Management, Handoff, Power Control And Link Adaptation InDistributed-Input Distributed-Output (DIDO) Communication Systems”

U.S. application Ser. No. 12/802,976, filed Jun. 16, 2010, entitled“System And Method For Adjusting DIDO Interference Cancellation Based OnSignal Strength Measurements”

U.S. application Ser. No. 12/802,974, filed Jun. 16, 2010, entitled“System And Method For Managing Inter-Cluster Handoff Of Clients WhichTraverse Multiple DIDO Clusters”

U.S. application Ser. No. 12/802,989, filed Jun. 16, 2010, entitled“System And Method For Managing Handoff Of A Client Between DifferentDistributed-Input-Distributed-Output (DIDO) Networks Based On DetectedVelocity Of The Client”

U.S. application Ser. No. 12/802,958, filed Jun. 16, 2010, entitled“System And Method For Power Control And Antenna Grouping In ADistributed-Input-Distributed-Output (DIDO) Network”

U.S. application Ser. No. 12/802,975, filed Jun. 16, 2010, entitled“System And Method For Link adaptation In DIDO Multicarrier Systems”

U.S. application Ser. No. 12/802,938, filed Jun. 16, 2010, entitled“System And Method For DIDO Precoding Interpolation In MulticarrierSystems”

U.S. application Ser. No. 12/630,627, filed Dec. 2, 2009, entitled“System and Method For Distributed Antenna Wireless Communications”

U.S. Pat. No. 7,599,420, filed Aug. 20, 2007, issued Oct. 6, 2009,entitled “System and Method for Distributed Input Distributed OutputWireless Communication”;

U.S. Pat. No. 7,633,994, filed Aug. 20, 2007, issued Dec. 15, 2009,entitled “System and Method for Distributed Input Distributed OutputWireless Communication”;

U.S. Pat. No. 7,636,381, filed Aug. 20, 2007, issued Dec. 22, 2009,entitled “System and Method for Distributed Input Distributed OutputWireless Communication”;

U.S. application Ser. No. 12/143,503, filed Jun. 20, 2008 entitled,“System and Method For Distributed Input-Distributed Output WirelessCommunications”;

U.S. application Ser. No. 11/256,478, filed Oct. 21, 2005 entitled“System and Method For Spatial-Multiplexed Tropospheric ScatterCommunications”;

U.S. Pat. No. 7,418,053, filed Jul. 30, 2004, issued Aug. 26, 2008,entitled “System and Method for Distributed Input Distributed OutputWireless Communication”;

U.S. application Ser. No. 10/817,731, filed Apr. 2, 2004 entitled“System and Method For Enhancing Near Vertical Incidence Skywave(“NVIS”) Communication Using Space-Time Coding.

To reduce the size and complexity of the present patent application, thedisclosure of some of the related patents and applications is notexplicitly set forth below. Please see the related patents andapplications for a full detailed description of the disclosure.

Note that section I below (Disclosure From Related application Ser. No.12/802,988) utilizes its own set of endnotes which refer to prior artreferences and prior applications assigned to the assignee of thepresent application. The endnote citations are listed at the end ofsection I (just prior to the heading for Section II). Citations inSection II uses may have numerical designations for its citations whichoverlap with those used in Section I even through these numericaldesignations identify different references (listed at the end of SectionII). Thus, references identified by a particular numerical designationmay be identified within the section in which the numerical designationis used.

I. Disclosure from Related Application Ser. No. 12/802,988

1. Methods to Remove Inter-Cluster Interference

Described below are wireless radio frequency (RF) communication systemsand methods employing a plurality of distributed transmitting antennasto create locations in space with zero RF energy. When M transmitantennas are employed, it is possible to create up to (M−1) points ofzero RF energy in predefined locations. In one embodiment of theinvention, the points of zero RF energy are wireless devices and thetransmit antennas are aware of the channel state information (CSI)between the transmitters and the receivers. In one embodiment, the CSIis computed at the receivers and fed back to the transmitters. Inanother embodiment, the CSI is computed at the transmitter via trainingfrom the receivers, assuming channel reciprocity is exploited. Thetransmitters may utilize the CSI to determine the interfering signals tobe simultaneously transmitted. In one embodiment, block diagonalization(BD) precoding is employed at the transmit antennas to generate pointsof zero RF energy.

The system and methods described herein differ from the conventionalreceive/transmit beamforming techniques described above. In fact,receive beamforming computes the weights to suppress interference at thereceive side (via null-steering), whereas some embodiments of theinvention described herein apply weights at the transmit side to createinterference patters that result in one or multiple locations in spacewith “zero RF energy.” Unlike conventional transmit beamforming or BDprecoding designed to maximize signal quality (or SINR) to every user ordownlink throughput, respectively, the systems and methods describedherein minimize signal quality under certain conditions and/or fromcertain transmitters, thereby creating points of zero RF energy at theclient devices (sometimes referred to herein as “users”). Moreover, inthe context of distributed-input distributed-output (DIDO) systems(described in our related patents and applications), transmit antennasdistributed in space provide higher degrees of freedom (i.e., higherchannel spatial diversity) that can be exploited to create multiplepoints of zero RF energy and/or maximum SINR to different users. Forexample, with M transmit antennas it is possible to create up to (M−1)points of RF energy. By contrast, practical beamforming or BD multiusersystems are typically designed with closely spaced antennas at thetransmit side that limit the number of simultaneous users that can beserviced over the wireless link, for any number of transmit antennas M.

Consider a system with M transmit antennas and K users, with K<M. Weassume the transmitter is aware of the CSI (H∈C^(K×M)) between the Mtransmit antennas and K users. For simplicity, every user is assumed tobe equipped with single antenna, but the same method can be extended tomultiple receive antennas per user. The precoding weights (w∈C^(M×1))that create zero RF energy at the K users' locations are computed tosatisfy the following conditionHw=0^(K×1)

-   -   where 0^(K×1) is the vector with all zero entries and H is the        channel matrix obtained by combining the channel vectors        (h_(k)∈C^(1×M)) from the M transmit antennas to the K users as

$H = {\begin{bmatrix}h_{1} \\\vdots \\h_{k} \\\vdots \\h_{K}\end{bmatrix}.}$

-   -   In one embodiment, singular value decomposition (SVD) of the        channel matrix H is computed and the precoding weight w is        defined as the right singular vector corresponding to the null        subspace (identified by zero singular value) of H.    -   The transmit antennas employ the weight vector defined above to        transmit RF energy, while creating K points of zero RF energy at        the locations of the K users such that the signal received at        the k^(th) user is given by        r _(k) =h _(k) ws _(k) +n _(k)=0+n _(k)    -   where n_(k)∈C^(1×1) is the additive white Gaussian noise (AWGN)        at the k^(th) user.        In one embodiment, singular value decomposition (SVD) of the        channel matrix H is computed and the precoding weight w is        defined as the right singular vector corresponding to the null        subspace (identified by zero singular value) of H.

In another embodiment, the wireless system is a DIDO system and pointsof zero RF energy are created to pre-cancel interference to the clientsbetween different DIDO coverage areas. In U.S. application Ser. No.12/630,627, a DIDO system is described which includes:

-   -   DIDO clients    -   DIDO distributed antennas    -   DIDO base transceiver stations (BTS)    -   DIDO base station network (BSN)        Every BTS is connected via the BSN to multiple distributed        antennas that provide service to given coverage area called DIDO        cluster. In the present patent application we describe a system        and method for removing interference between adjacent DIDO        clusters. As illustrated in FIG. 1, we assume the main DIDO        cluster hosts the client (i.e. a user device served by the        multi-user DIDO system) affected by interference (or target        client) from the neighbor clusters.

In one embodiment, neighboring clusters operate at different frequenciesaccording to frequency division multiple access (FDMA) techniquessimilar to conventional cellular systems. For example, with frequencyreuse factor of 3, the same carrier frequency is reused every third DIDOcluster as illustrated in FIG. 2. In FIG. 2, the different carrierfrequencies are identified as F₁, F₂ and F₃. While this embodiment maybe used in some implementations, this solution yields loss in spectralefficiency since the available spectrum is divided in multiple subbandsand only a subset of DIDO clusters operate in the same subband.Moreover, it requires complex cell planning to associate different DIDOclusters to different frequencies, thereby preventing interference. Likeprior art cellular systems, such cellular planning requires specificplacement of antennas and limiting of transmit power to as to avoidinterference between clusters using the same frequency.

In another embodiment, neighbor clusters operate in the same frequencyband, but at different time slots according to time division multipleaccess (TDMA) technique. For example, as illustrated in FIG. 3 DIDOtransmission is allowed only in time slots T₁, T₂, and T₃ for certainclusters, as illustrated. Time slots can be assigned equally todifferent clusters, such that different clusters are scheduled accordingto a Round-Robin policy. If different clusters are characterized bydifferent data rate requirements (i.e., clusters in crowded urbanenvironments as opposed to clusters in rural areas with fewer number ofclients per area of coverage), different priorities are assigned todifferent clusters such that more time slots are assigned to theclusters with larger data rate requirements. While TDMA as describedabove may be employed in one embodiment of the invention, a TDMAapproach may require time synchronization across different clusters andmay result in lower spectral efficiency since interfering clusterscannot use the same frequency at the same time.

In one embodiment, all neighboring clusters transmit at the same time inthe same frequency band and use spatial processing across clusters toavoid interference. In this embodiment, the multi-cluster DIDO system:(i) uses conventional DIDO precoding within the main cluster to transmitsimultaneous non-interfering data streams within the same frequency bandto multiple clients (such as described in the related patents andapplications, including U.S. Pat. Nos. 7,599,420; 7,633,994; 7,636,381;and application Ser. No. 12/143,503); (ii) uses DIDO precoding withinterference cancellation in the neighbor clusters to avoid interferenceto the clients lying in the interfering zones 8010 in FIG. 4, bycreating points of zero radio frequency (RF) energy at the locations ofthe target clients. If a target client is in an interfering zone 410, itwill receive the sum of the RF containing the data stream from the maincluster 411 and the zero RF energy from the interfering cluster 412-413,which will simply be the RF containing the data stream from the maincluster. Thus, adjacent clusters can utilize the same frequencysimultaneously without target clients in the interfering zone sufferingfrom interference.

In practical systems, the performance of DIDO precoding may be affectedby different factors such as: channel estimation error or Dopplereffects (yielding obsolete channel state information at the DIDOdistributed antennas); intermodulation distortion (IMD) in multicarrierDIDO systems; time or frequency offsets. As a result of these effects,it may be impractical to achieve points of zero RF energy. However, aslong as the RF energy at the target client from the interfering clustersis negligible compared to the RF energy from the main cluster, the linkperformance at the target client is unaffected by the interference. Forexample, let us assume the client requires 20 dB signal-to-noise ratio(SNR) to demodulate 4-QAM constellations using forward error correction(FEC) coding to achieve target bit error rate (BER) of 10⁻⁶. If the RFenergy at the target client received from the interfering cluster is 20dB below the RF energy received from the main cluster, the interferenceis negligible and the client can demodulate data successfully within thepredefined BER target. Thus, the term “zero RF energy” as used hereindoes not necessarily mean that the RF energy from interfering RF signalsis zero. Rather, it means that the RF energy is sufficiently lowrelative to the RF energy of the desired RF signal such that the desiredRF signal may be received at the receiver. Moreover, while certaindesirable thresholds for interfering RF energy relative to desired RFenergy are described, the underlying principles of the invention are notlimited to any particular threshold values.

There are different types of interfering zones 8010 as shown in FIG. 4.For example, “type A” zones (as indicated by the letter “A” in FIG. 80)are affected by interference from only one neighbor cluster, whereas“type B” zones (as indicated by the letter “B”) account for interferencefrom two or multiple neighbor clusters.

FIG. 5 depicts a framework employed in one embodiment of the invention.The dots denote DIDO distributed antennas, the crosses refer to the DIDOclients and the arrows indicate the directions of propagation of RFenergy. The DIDO antennas in the main cluster transmit precoded datasignals to the clients MC 501 in that cluster. Likewise, the DIDOantennas in the interfering cluster serve the clients IC 502 within thatcluster via conventional DIDO precoding. The green cross 503 denotes thetarget client TC 503 in the interfering zone. The DIDO antennas in themain cluster 511 transmit precoded data signals to the target client(black arrows) via conventional DIDO precoding. The DIDO antennas in theinterfering cluster 512 use precoding to create zero RF energy towardsthe directions of the target client 503 (green arrows).

The received signal at target client k in any interfering zone 410A, Bin FIG. 4 is given by

$\begin{matrix}{r_{k} = {{H_{k}W_{k}s_{k}} + {H_{k}\underset{u \neq k}{\Sigma_{u = 1}^{U}}W_{u}s_{u}} + {\Sigma_{c = 1}^{C}H_{c,k}\Sigma_{i = 1}^{I_{c}}W_{c,i}s_{c,i}} + n_{k}}} & (1)\end{matrix}$where k=1, . . . , K, with K being the number of clients in theinterfering zone 8010A, B, U is the number of clients in the main DIDOcluster, C is the number of interfering DIDO clusters 412-413 and I_(c)is the number of clients in the interfering cluster c. Moreover,r_(k)∈C^(N×M) is the vector containing the receive data streams atclient k, assuming M transmit DIDO antennas and N receive antennas atthe client devices; s_(k)∈C^(N×1) is the vector of transmit data streamsto client k in the main DIDO cluster; s_(u)∈C^(N×1) is the vector oftransmit data streams to client u in the main DIDO cluster;s_(c,i)∈C^(N×1) is the vector of transmit data streams to client i inthe c^(th) interfering DIDO cluster; n_(k)∈C^(N×1) is the vector ofadditive white Gaussian noise (AWGN) at the N receive antennas of clientk; H_(k)∈C^(N×M) is the DIDO channel matrix from the M transmit DIDOantennas to the N receive antennas at client k in the main DIDO cluster;H_(c,k)∈C^(N×M) is the DIDO channel matrix from the M transmit DIDOantennas to the N receive antennas t client k in the c^(th) interferingDIDO cluster; W_(k)∈C^(M×N) is the matrix of DIDO precoding weights toclient k in the main DIDO cluster; W_(k)∈C^(M×N) is the matrix of DIDOprecoding weights to client u in the main DIDO cluster; W_(c,i)∈C^(M×N)is the matrix of DIDO precoding weights to client i in the c^(th)interfering DIDO cluster.

To simplify the notation and without loss of generality, we assume allclients are equipped with N receive antennas and there are M DIDOdistributed antennas in every DIDO cluster, with M≥(N·U) andM≥(N·I_(c)), ∀c=1, . . . , C. If M is larger than the total number ofreceive antennas in the cluster, the extra transmit antennas are used topre-cancel interference to the target clients in the interfering zone orto improve link robustness to the clients within the same cluster viadiversity schemes described in the related patents and applications,including U.S. Pat. Nos. 7,599,420; 7,633,994; 7,636,381; andapplication Ser. No. 12/143,503.

The DIDO precoding weights are computed to pre-cancel inter-clientinterference within the same DIDO cluster. For example, blockdiagonalization (BD) precoding described in the related patents andapplications, including U.S. Pat. Nos. 7,599,420; 7,633,994; 7,636,381;and application Ser. No. 12/143,503 and [7] can be used to removeinter-client interference, such that the following condition issatisfied in the main clusterH _(k) W _(u)=0^(N×N) ; ∀u=1, . . . ,U; with u≠k.  (2)The precoding weight matrices in the neighbor DIDO clusters are designedsuch that the following condition is satisfiedH _(c,k) W _(c,i)=0^(N×N) ; ∀c=1, . . . ,C and ∀i=1, . . . ,I _(c).  (3)To compute the precoding matrices W_(c,i), the downlink channel from theM transmit antennas to the I_(c) clients in the interfering cluster aswell as to client k in the interfering zone is estimated and theprecoding matrix is computed by the DIDO BTS in the interfering cluster.If BD method is used to compute the precoding matrices in theinterfering clusters, the following effective channel matrix is built tocompute the weights to the i^(th) client in the neighbor clusters

$\begin{matrix}{{\overset{\_}{H}}_{c,i} = \begin{bmatrix}H_{c,k} \\{\overset{\sim}{H}}_{c,i}\end{bmatrix}} & (4)\end{matrix}$where {tilde over (H)}_(c,i) is the matrix obtained from the channelmatrix H_(c)∈C^((N·I) ^(c) ^()×M) for the interfering cluster c, wherethe rows corresponding to the i^(th) client are removed.

Substituting conditions (2) and (3) into (1), we obtain the receiveddata streams for target client k, where intra-cluster and inter-clusterinterference is removedr _(k) =H _(k) W _(k) s _(k) +n _(k).  (5)The precoding weights W_(c,i) in (1) computed in the neighbor clustersare designed to transmit precoded data streams to all clients in thoseclusters, while pre-cancelling interference to the target client in theinterfering zone. The target client receives precoded data only from itsmain cluster. In a different embodiment, the same data stream is sent tothe target client from both main and neighbor clusters to obtaindiversity gain. In this case, the signal model in (5) is expressed asr _(k)=(H _(k) W _(k)+Σ_(c=1) ^(c) H _(c,k) W _(c,k))s _(k) +n _(k)  (6)where W_(c,k) is the DIDO precoding matrix from the DIDO transmitters inthe c^(th) cluster to the target client k in the interfering zone. Notethat the method in (6) requires time synchronization across neighboringclusters, which may be complex to achieve in large systems, butnonetheless, is quite feasible if the diversity gain benefit justifiesthe cost of implementation.

We begin by evaluating the performance of the proposed method in termsof symbol error rate (SER) as a function of the signal-to-noise ratio(SNR). Without loss of generality, we define the following signal modelassuming single antenna per client and reformulate (1) asr _(k)=√{square root over (SNR)}h _(k) w _(k) s _(k)+√{square root over(INR)}h _(c,k)Σ_(i=1) ^(I) w _(c,i) s _(c,i) +n _(k)  (7)where INR is the interference-to-noise ratio defined as INR=SNR/SIR andSIR is the signal-to-interference ratio.

FIG. 6 shows the SER as a function of the SNR, assuming SIR=10 dB forthe target client in the interfering zone. Without loss of generality,we measured the SER for 4-QAM and 16-QAM without forwards errorcorrection (FEC) coding. We fix the target SER to 1% for uncodedsystems. This target corresponds to different values of SNR depending onthe modulation order (i.e., SNR=20 dB for 4-QAM and SNR=28 dB for16-QAM). Lower SER targets can be satisfied for the same values of SNRwhen using FEC coding due to coding gain. We consider the scenario oftwo clusters (one main cluster and one interfering cluster) with twoDIDO antennas and two clients (equipped with single antenna each) percluster. One of the clients in the main cluster lies in the interferingzone. We assume flat-fading narrowband channels, but the followingresults can be extended to frequency selective multicarrier (OFDM)systems, where each subcarrier undergoes flat-fading. We consider twoscenarios: (i) one with inter-DIDO-cluster interference (IDCI) where theprecoding weights w_(c,i) are computed without accounting for the targetclient in the interfering zone; and (ii) the other where the IDCI isremoved by computing the weights w_(c,i) to cancel IDCI to the targetclient. We observe that in presence of IDCI the SER is high and abovethe predefined target. With IDCI-precoding at the neighbor cluster theinterference to the target client is removed and the SER targets arereached for SNR>20 dB.

The results in FIG. 6 assumes IDCI-precoding as in (5). IfIDCI-precoding at the neighbor clusters is also used to precode datastreams to the target client in the interfering zone as in (6),additional diversity gain is obtained. FIG. 7 compares the SER derivedfrom two techniques: (i) “Method 1” using the IDCI-precoding in (5);(ii) “Method 2” employing IDCI-precoding in (6) where the neighborclusters also transmit precoded data stream to the target client. Method2 yields ˜3 dB gain compared to conventional IDCI-precoding due toadditional array gain provided by the DIDO antennas in the neighborcluster used to transmit precoded data stream to the target client. Moregenerally, the array gain of Method 2 over Method 1 is proportional to10*log 10(C+1), where C is the number of neighbor clusters and thefactor “1” refers to the main cluster.

Next, we evaluate the performance of the above method as a function ofthe target client's location with respect to the interfering zone. Weconsider one simple scenario where a target client 8401 moves from themain DIDO cluster 802 to the interfering cluster 803, as depicted inFIG. 8. We assume all DIDO antennas 812 within the main cluster 802employ BD precoding to cancel intra-cluster interference to satisfycondition (2). We assume single interfering DIDO cluster, singlereceiver antenna at the client device 801 and equal pathloss from allDIDO antennas in the main or interfering cluster to the client (i.e.,DIDO antennas placed in circle around the client). We use one simplifiedpathloss model with pathloss exponent 4 (as in typical urbanenvironments) [11].

The analysis hereafter is based on the following simplified signal modelthat extends (7) to account for pathloss

$\begin{matrix}{r_{k} = {{\sqrt{\frac{{SNR} \cdot D_{o}^{4}}{D^{4}}}h_{k}w_{k}s_{k}} + {\sqrt{\frac{{SNR} \cdot D_{o}^{4}}{( {1 - D} )^{4}}}h_{c,k}\Sigma_{i = 1}^{I}w_{c,i}s_{c,i}} + n_{k}}} & (8)\end{matrix}$where the signal-to-interference (SIR) is derived as SIR=((1−D)/D)⁴. Inmodeling the IDCI, we consider three scenarios: i) ideal case with noIDCI; ii) IDCI pre-cancelled via BD precoding in the interfering clusterto satisfy condition (3); iii) with IDCI, not pre-cancelled by theneighbor cluster.

FIG. 9 shows the signal-to-interference-plus-noise ratio (SINR) as afunction of D (i.e., as the target client moves from the main cluster802 towards the DIDO antennas 813 in the interfering cluster 8403). TheSINR is derived as the ratio of signal power and interference plus noisepower using the signal model in (8). We assume that D_(o)=0.1 and SNR=50dB for D=D_(o). In absence of IDCI the wireless link performance is onlyaffected by noise and the SINR decreases due to pathloss. In presence ofIDCI (i.e., without IDCI-precoding) the interference from the DIDOantennas in the neighbor cluster contributes to reduce the SINR.

FIG. 10 shows the symbol error rate (SER) performance of the threescenarios above for 4-QAM modulation in flat-fading narrowband channels.These SER results correspond to the SINR in FIG. 9. We assume SERthreshold of 1% for uncoded systems (i.e., without FEC) corresponding toSINR threshold SINR_(T)=20 dB in FIG. 9. The SINR threshold depends onthe modulation order used for data transmission. Higher modulationorders are typically characterized by higher SINR_(T) to achieve thesame target error rate. With FEC, lower target SER can be achieved forthe same SINR value due to coding gain. In case of IDCI withoutprecoding, the target SER is achieved only within the range D<0.25. WithIDCI-precoding at the neighbor cluster the range that satisfies thetarget SER is extended up to D<0.6. Beyond that range, the SINRincreases due to pathloss and the SER target is not satisfied.

One embodiment of a method for IDCI precoding is shown in FIG. 11 andconsists of the following steps:

-   -   SIR estimate 1101: Clients estimate the signal power from the        main DIDO cluster (i.e., based on received precoded data) and        the interference-plus-noise signal power from the neighbor DIDO        clusters. In single-carrier DIDO systems, the frame structure        can be designed with short periods of silence. For example,        periods of silence can be defined between training for channel        estimation and precoded data transmissions during channel state        information (CSI) feedback. In one embodiment, the        interference-plus-noise signal power from neighbor clusters is        measured during the periods of silence from the DIDO antennas in        the main cluster. In practical DIDO multicarrier (OFDM) systems,        null tones are typically used to prevent direct current (DC)        offset and attenuation at the edge of the band due to filtering        at transmit and receive sides. In another embodiment employing        multicarrier systems, the interference-plus-noise signal power        is estimated from the null tones. Correction factors can be used        to compensate for transmit/receive filter attenuation at the        edge of the band. Once the signal-plus-interference-and-noise        power (P_(S)) from the main cluster and the        interference-plus-noise power from neighbor clusters (P_(IN))        are estimated, the client computes the SINR as

$\begin{matrix}{{SINR} = {\frac{P_{S} - P_{IN}}{P_{IN}}.}} & (9)\end{matrix}$

-   -   Alternatively, the SINR estimate is derived from the received        signal strength indication (RSSI) used in typical wireless        communication systems to measure the radio signal power.    -   We observe the metric in (9) cannot discriminate between noise        and interference power level. For example, clients affected by        shadowing (i.e., behind obstacles that attenuate the signal        power from all DIDO distributed antennas in the main cluster) in        interference-free environments may estimate low SINR even though        they are not affected by inter-cluster interference. A more        reliable metric for the proposed method is the SIR computed as

$\begin{matrix}{{SIR} = \frac{P_{S} - P_{IN}}{P_{IN} - P_{N}}} & (10)\end{matrix}$

-   -   where P_(N) is the noise power. In practical multicarrier OFDM        systems, the noise power P_(N) in (10) is estimated from the        null tones, assuming all DIDO antennas from main and neighbor        clusters use the same set of null tones. The        interference-plus-noise power (P_(IN)), is estimated from the        period of silence as mentioned above. Finally, the        signal-plus-interference-and-noise power (P_(S)) is derived from        the data tones. From these estimates, the client computes the        SIR in (10).    -   Channel estimation at neighbor clusters 1102-1103: If the        estimated SIR in (10) is below predefined threshold (SIR_(T)),        determined at 8702 in FIG. 11, the client starts listening to        training signals from neighbor clusters. Note that SIR_(T)        depends on the modulation and FEC coding scheme (MCS) used for        data transmission. Different SIR targets are defined depending        on the client's MCS. When DIDO distributed antennas from        different clusters are time-synchronized (i.e., locked to the        same pulse-per-second, PPS, time reference), the client exploits        the training sequence to deliver its channel estimates to the        DIDO antennas in the neighbor clusters at 8703. The training        sequence for channel estimation in the neighbor clusters are        designed to be orthogonal to the training from the main cluster.        Alternatively, when DIDO antennas in different clusters are not        time-synchronized, orthogonal sequences (with good        cross-correlation properties) are used for time synchronization        in different DIDO clusters. Once the client locks to the        time/frequency reference of the neighbor clusters, channel        estimation is carried out at 1103.    -   IDCI Precoding 1104: Once the channel estimates are available at        the DIDO BTS in the neighbor clusters, IDCI-precoding is        computed to satisfy the condition in (3). The DIDO antennas in        the neighbor clusters transmit precoded data streams only to the        clients in their cluster, while pre-cancelling interference to        the clients in the interfering zone 410 in FIG. 4. We observe        that if the client lies in the type B interfering zone 410 in        FIG. 4, interference to the client is generated by multiple        clusters and IDCI-precoding is carried out by all neighbor        clusters at the same time.        Methods for Handoff

Hereafter, we describe different handoff methods for clients that moveacross DIDO clusters populated by distributed antennas that are locatedin separate areas or that provide different kinds of services (i.e.,low- or high-mobility services).

a. Handoff Between Adjacent DIDO Clusters

In one embodiment, the IDCI-precoder to remove inter-clusterinterference described above is used as a baseline for handoff methodsin DIDO systems. Conventional handoff in cellular systems is conceivedfor clients to switch seamlessly across cells served by different basestations. In DIDO systems, handoff allows clients to move from onecluster to another without loss of connection.

To illustrate one embodiment of a handoff strategy for DIDO systems, weconsider again the example in FIG. 8 with only two clusters 802 and 803.As the client 801 moves from the main cluster (C1) 802 to the neighborcluster (C2) 803, one embodiment of a handoff method dynamicallycalculates the signal quality in different clusters and selects thecluster that yields the lowest error rate performance to the client.

FIG. 12 shows the SINR variation as a function of the client's distancefrom the center of clusters C1. For 4-QAM modulation without FEC coding,we consider target SINR=20 dB. The line identified by circles representsthe SINR for the target client being served by the DIDO antennas in C1,when both C1 and C2 use DIDO precoding without interferencecancellation. The SINR decreases as a function of D due to pathloss andinterference from the neighboring cluster. When IDCI-precoding isimplemented at the neighboring cluster, the SINR loss is only due topathloss (as shown by the line with triangles), since interference iscompletely removed. Symmetric behavior is experienced when the client isserved from the neighboring cluster. One embodiment of the handoffstrategy is defined such that, as the client moves from C1 to C2, thealgorithm switches between different DIDO schemes to maintain the SINRabove predefined target.

From the plots in FIG. 12, we derive the SER for 4-QAM modulation inFIG. 13. We observe that, by switching between different precodingstrategies, the SER is maintained within predefined target.

One embodiment of the handoff strategy is as follows.

-   -   C1-DIDO and C2-DIDO precoding: When the client lies within C1,        away from the interfering zone, both clusters C1 and C2 operate        with conventional DIDO precoding independently.    -   C1-DIDO and C2-IDCI precoding: As the client moves towards the        interfering zone, its SIR or SINR degrades. When the target        SINR_(T1) is reached, the target client starts estimating the        channel from all DIDO antennas in C2 and provides the CSI to the        BTS of C2. The BTS in C2 computes IDCI-precoding and transmits        to all clients in C2 while preventing interference to the target        client. For as long as the target client is within the        interfering zone, it will continue to provide its CSI to both C1        and C2.    -   C1-IDCI and C2-DIDO precoding: As the client moves towards C2,        its SIR or SINR keeps decreasing until it again reaches a        target. At this point the client decides to switch to the        neighbor cluster. In this case, C1 starts using the CSI from the        target client to create zero interference towards its direction        with IDCI-precoding, whereas the neighbor cluster uses the CSI        for conventional DIDO-precoding. In one embodiment, as the SIR        estimate approaches the target, the clusters C1 and C2 try both        DIDO- and IDCI-precoding schemes alternatively, to allow the        client to estimate the SIR in both cases. Then the client        selects the best scheme, to maximize certain error rate        performance metric. When this method is applied, the cross-over        point for the handoff strategy occurs at the intersection of the        curves with triangles and rhombus in FIG. 12. One embodiment        uses the modified IDCI-precoding method described in (6) where        the neighbor cluster also transmits precoded data stream to the        target client to provide array gain. With this approach the        handoff strategy is simplified, since the client does not need        to estimate the SINR for both strategies at the cross-over        point.    -   C1-DIDO and C2-DIDO precoding: As the client moves out of the        interference zone towards C2, the main cluster C1 stops        pre-cancelling interference towards that target client via        IDCI-precoding and switches back to conventional DIDO-precoding        to all clients remaining in C1. This final cross-over point in        our handoff strategy is useful to avoid unnecessary CSI feedback        from the target client to C1, thereby reducing the overhead over        the feedback channel. In one embodiment a second target        SINR_(T2) is defined. When the SINR (or SIR) increases above        this target, the strategy is switched to C1-DIDO and C2-DIDO. In        one embodiment, the cluster C1 keeps alternating between DIDO-        and IDCI-precoding to allow the client to estimate the SINR.        Then the client selects the method for C1 that more closely        approaches the target SINR_(T1) from above.

The method described above computes the SINR or SIR estimates fordifferent schemes in real time and uses them to select the optimalscheme. In one embodiment, the handoff algorithm is designed based onthe finite-state machine illustrated in FIG. 14. The client keeps trackof its current state and switches to the next state when the SINR or SIRdrops below or above the predefined thresholds illustrated in FIG. 12.As discussed above, in state 1201, both clusters C1 and C2 operate withconventional DIDO precoding independently and the client is served bycluster C1; in state 1202, the client is served by cluster C1, the BTSin C2 computes IDCI-precoding and cluster C1 operates using conventionalDIDO precoding; in state 1203, the client is served by cluster C2, theBTS in C1 computes IDCI-precoding and cluster C2 operates usingconventional DIDO precoding; and in state 1204, the client is served bycluster C2, and both clusters C1 and C2 operate with conventional DIDOprecoding independently.

In presence of shadowing effects, the signal quality or SIR mayfluctuate around the thresholds as shown in FIG. 15, causing repetitiveswitching between consecutive states in FIG. 14. Changing statesrepetitively is an undesired effect, since it results in significantoverhead on the control channels between clients and BTSs to enableswitching between transmission schemes. FIG. 15 depicts one example of ahandoff strategy in the presence of shadowing. In one embodiment, theshadowing coefficient is simulated according to log-normal distributionwith variance 3 [3]. Hereafter, we define some methods to preventrepetitive switching effect during DIDO handoff.

One embodiment of the invention employs a hysteresis loop to cope withstate switching effects. For example, when switching between“C1-DIDO,C2-IDCI” 9302 and “C1-IDCI,C2-DIDO” 9303 states in FIG. 14 (orvice versa) the threshold SINR_(T1) can be adjusted within the range A₁.This method avoids repetitive switches between states as the signalquality oscillates around SINR_(T1). For example, FIG. 16 shows thehysteresis loop mechanism when switching between any two states in FIG.14. To switch from state B to A the SIR must be larger than(SIR_(T1)+A₁/2), but to switch back from A to B the SIR must drop below(SIR_(T1)−A₁/2).

In a different embodiment, the threshold SINR_(T2) is adjusted to avoidrepetitive switching between the first and second (or third and fourth)states of the finite-state machine in FIG. 14. For example, a range ofvalues A₂ may be defined such that the threshold SINR_(T2) is chosenwithin that range depending on channel condition and shadowing effects.

In one embodiment, depending on the variance of shadowing expected overthe wireless link, the SINR threshold is dynamically adjusted within therange [SINR_(T2), SINR_(T2)+A₂]. The variance of the log-normaldistribution can be estimated from the variance of the received signalstrength (or RSSI) as the client moves from its current cluster to theneighbor cluster.

The methods above assume the client triggers the handoff strategy. Inone embodiment, the handoff decision is deferred to the DIDO BTSs,assuming communication across multiple BTSs is enabled.

For simplicity, the methods above are derived assuming no FEC coding and4-QAM. More generally, the SINR or SIR thresholds are derived fordifferent modulation coding schemes (MCSs) and the handoff strategy isdesigned in combination with link adaptation (see, e.g., U.S. Pat. No.7,636,381) to optimize downlink data rate to each client in theinterfering zone.

b. Handoff Between Low- and High-Doppler DIDO Networks

DIDO systems employ closed-loop transmission schemes to precode datastreams over the downlink channel. Closed-loop schemes are inherentlyconstrained by latency over the feedback channel. In practical DIDOsystems, computational time can be reduced by transceivers with highprocessing power and it is expected that most of the latency isintroduced by the DIDO BSN, when delivering CSI and baseband precodeddata from the BTS to the distributed antennas. The BSN can be comprisedof various network technologies including, but not limited to, digitalsubscriber lines (DSL), cable modems, fiber rings, T1 lines, hybridfiber coaxial (HFC) networks, and/or fixed wireless (e.g., WiFi).Dedicated fiber typically has very large bandwidth and low latency,potentially less than a millisecond in local region, but it is lesswidely deployed than DSL and cable modems. Today, DSL and cable modemconnections typically have between 10-25 ms in last-mile latency in theUnited States, but they are very widely deployed.

The maximum latency over the BSN determines the maximum Dopplerfrequency that can be tolerated over the DIDO wireless link withoutperformance degradation of DIDO precoding. For example, in [1] we showedthat at the carrier frequency of 400 MHz, networks with latency of about10 msec (i.e., DSL) can tolerate clients' velocity up to 8 mph (runningspeed), whereas networks with 1 msec latency (i.e., fiber ring) cansupport speed up to 70 mph (i.e., freeway traffic).

We define two or multiple DIDO sub-networks depending on the maximumDoppler frequency that can be tolerated over the BSN. For example, a BSNwith high-latency DSL connections between the DIDO BTS and distributedantennas can only deliver low mobility or fixed-wireless services (i.e.,low-Doppler network), whereas a low-latency BSN over a low-latency fiberring can tolerate high mobility (i.e., high-Doppler network). We observethat the majority of broadband users are not moving when they usebroadband, and further, most are unlikely to be located near areas withmany high speed objects moving by (e.g., next to a highway) since suchlocations are typically less desirable places to live or operate anoffice. However, there are broadband users who will be using broadbandat high speeds (e.g., while in a car driving on the highway) or will benear high speed objects (e.g., in a store located near a highway). Toaddress these two differing user Doppler scenarios, in one embodiment, alow-Doppler DIDO network consists of a typically larger number of DIDOantennas with relatively low power (i.e., 1 W to 100 W, for indoor orrooftop installation) spread across a wide area, whereas a high-Dopplernetwork consists of a typically lower number of DIDO antennas with highpower transmission (i.e., 100 W for rooftop or tower installation). Thelow-Doppler DIDO network serves the typically larger number oflow-Doppler users and can do so at typically lower connectivity costusing inexpensive high-latency broadband connections, such as DSL andcable modems. The high-Doppler DIDO network serves the typically fewernumber of high-Doppler users and can do so at typically higherconnectivity cost using more expensive low-latency broadbandconnections, such as fiber.

To avoid interference across different types of DIDO networks (e.g.low-Doppler and high-Doppler), different multiple access techniques canbe employed such as: time division multiple access (TDMA), frequencydivision multiple access (FDMA), or code division multiple access(CDMA).

Hereafter, we propose methods to assign clients to different types ofDIDO networks and enable handoff between them. The network selection isbased on the type of mobility of each client. The client's velocity (v)is proportional to the maximum Doppler shift according to the followingequation [6]

$\begin{matrix}{f_{d} = {\frac{v}{\lambda}\sin\;\theta}} & (11)\end{matrix}$where f_(d) is the maximum Doppler shift, λ is the wavelengthcorresponding to the carrier frequency and θ is the angle between thevector indicating the direction transmitter-client and the velocityvector.

In one embodiment, the Doppler shift of every client is calculated viablind estimation techniques. For example, the Doppler shift can beestimated by sending RF energy to the client and analyzing the reflectedsignal, similar to Doppler radar systems.

In another embodiment, one or multiple DIDO antennas send trainingsignals to the client. Based on those training signals, the clientestimates the Doppler shift using techniques such as counting thezero-crossing rate of the channel gain, or performing spectrum analysis.We observe that for fixed velocity v and client's trajectory, theangular velocity v sin θ in (11) may depend on the relative distance ofthe client from every DIDO antenna. For example, DIDO antennas in theproximity of a moving client yield larger angular velocity and Dopplershift than faraway antennas. In one embodiment, the Doppler velocity isestimated from multiple DIDO antennas at different distances from theclient and the average, weighted average or standard deviation is usedas an indicator for the client's mobility. Based on the estimatedDoppler indicator, the DIDO BTS decides whether to assign the client tolow- or high-Doppler networks.

The Doppler indicator is periodically monitored for all clients and sentback to the BTS. When one or multiple clients change their Dopplervelocity (i.e., client riding in the bus versus client walking orsitting), those clients are dynamically re-assigned to different DIDOnetwork that can tolerate their level of mobility.

Although the Doppler of low-velocity clients can be affected by being inthe vicinity of high-velocity objects (e.g. near a highway), the Doppleris typically far less than the Doppler of clients that are in motionthemselves. As such, in one embodiment, the velocity of the client isestimated (e.g. by using a means such as monitoring the clients positionusing GPS), and if the velocity is low, the client is assigned to alow-Doppler network, and if the velocity if high, the client is assignedto a high-Doppler network.

Methods for Power Control and Antenna Grouping

The block diagram of DIDO systems with power control is depicted in FIG.17. One or multiple data streams (s_(k)) for every client (1, . . . , U)are first multiplied by the weights generated by the DIDO precodingunit. Precoded data streams are multiplied by power scaling factorcomputed by the power control unit, based on the input channel qualityinformation (CQI). The CQI is either fed back from the clients to DIDOBTS or derived from the uplink channel assuming uplink-downlink channelreciprocity. The U precoded streams for different clients are thencombined and multiplexed into M data streams (t_(m)), one for each ofthe M transmit antennas. Finally, the streams t_(m) are sent to thedigital-to-analog converter (DAC) unit, the radio frequency (RF) unit,power amplifier (PA) unit and finally to the antennas.

The power control unit measures the CQI for all clients. In oneembodiment, the CQI is the average SNR or RSSI. The CQI varies fordifferent clients depending on pathloss or shadowing. Our power controlmethod adjusts the transmit power scaling factors P_(k) for differentclients and multiplies them by the precoded data streams generated fordifferent clients. Note that one or multiple data streams may begenerated for every client, depending on the number of clients' receiveantennas.

To evaluate the performance of the proposed method, we defined thefollowing signal model based on (5), including pathloss and powercontrol parametersr _(k)=√{square root over (SNRP _(k)α_(k))}H _(k) W _(k) s _(k) +n_(k)  (12)where k=1, . . . , U, U is the number of clients, SNR=P_(o)/N_(o), withP_(o) being the average transmit power, N_(o) the noise power and α_(k)the pathloss/shadowing coefficient. To model pathloss/shadowing, we usethe following simplified model

$\begin{matrix}{\alpha_{k} = e^{{- a}\;\frac{k - 1}{U}}} & (13)\end{matrix}$where a=4 is the pathloss exponent and we assume the pathloss increaseswith the clients' index (i.e., clients are located at increasingdistance from the DIDO antennas).

FIG. 18 shows the SER versus SNR assuming four DIDO transmit antennasand four clients in different scenarios. The ideal case assumes allclients have the same pathloss (i.e., a=0), yielding P_(k)=1 for allclients. The plot with squares refers to the case where clients havedifferent pathloss coefficients and no power control. The curve withdots is derived from the same scenario (with pathloss) where the powercontrol coefficients are chosen such that P_(k)=1/α_(k). With the powercontrol method, more power is assigned to the data streams intended tothe clients that undergo higher pathloss/shadowing, resulting in 9 dBSNR gain (for this particular scenario) compared to the case with nopower control.

The Federal Communications Commission (FCC) (and other internationalregulatory agencies) defines constraints on the maximum power that canbe transmitted from wireless devices to limit the exposure of human bodyto electromagnetic (EM) radiation. There are two types of limits [2]: i)“occupational/controlled” limit, where people are made fully aware ofthe radio frequency (RF) source via fences, warnings or labels; ii)“general population/uncontrolled” limit where there is no control overthe exposure.

Different emission levels are defined for different types of wirelessdevices. In general, DIDO distributed antennas used for indoor/outdoorapplications qualify for the FCC category of “mobile” devices, definedas [2]:

“transmitting devices designed to be used in other than fixed locationsthat would normally be used with radiating structures maintained 20 cmor more from the body of the user or nearby persons.”

The EM emission of “mobile” devices is measured in terms of maximumpermissible exposure (MPE), expressed in mW/cm². FIG. 19 shows the MPEpower density as a function of distance from the source of RF radiationfor different values of transmit power at 700 MHz carrier frequency. Themaximum allowed transmit power to meet the FCC “uncontrolled” limit fordevices that typically operate beyond 20 cm from the human body is 1 W.

Less restrictive power emission constraints are defined for transmittersinstalled on rooftops or buildings, away from the “general population”.For these “rooftop transmitters” the FCC defines a looser emission limitof 1000 W, measured in terms of effective radiated power (ERP).

Based on the above FCC constraints, in one embodiment we define twotypes of DIDO distributed antennas for practical systems:

-   -   Low-power (LP) transmitters: located anywhere (i.e., indoor or        outdoor) at any height, with maximum transmit power of 1 W and 5        Mbps consumer-grade broadband (e.g. DSL, cable modem, Fibe To        The Home (FTTH)) backhaul connectivity.    -   High-power (HP) transmitters: rooftop or building mounted        antennas at height of approximately 10 meters, with transmit        power of 100 W and a commercial-grade broadband (e.g. optical        fiber ring) backhaul (with effectively “unlimited” data rate        compared to the throughput available over the DIDO wireless        links).

Note that LP transmitters with DSL or cable modem connectivity are goodcandidates for low-Doppler DIDO networks (as described in the previoussection), since their clients are mostly fixed or have low mobility. HPtransmitters with commercial fiber connectivity can tolerate higherclient's mobility and can be used in high-Doppler DIDO networks.

To gain practical intuition on the performance of DIDO systems withdifferent types of LP/HP transmitters, we consider the practical case ofDIDO antenna installation in downtown Palo Alto, Calif. FIG. 20a shows arandom distribution of N_(LP)=100 low-power DIDO distributed antennas inPalo Alto. In FIG. 20 b, 50 LP antennas are substituted with N_(HP)=50high-power transmitters.

Based on the DIDO antenna distributions in FIGS. 20a-b , we derive thecoverage maps in Palo Alto for systems using DIDO technology. FIGS. 21aand 21b show two power distributions corresponding to the configurationsin FIG. 20a and FIG. 20b , respectively. The received power distribution(expressed in dBm) is derived assuming the pathloss/shadowing model forurban environments defined by the 3GPP standard [3] at the carrierfrequency of 700 MHz. We observe that using 50% of HP transmittersyields better coverage over the selected area.

FIGS. 22a-b depict the rate distribution for the two scenarios above.The throughput (expressed in Mbps) is derived based on power thresholdsfor different modulation coding schemes defined in the 3GPP long-termevolution (LTE) standard in [4,5]. The total available bandwidth isfixed to 10 MHz at 700 MHz carrier frequency. Two different frequencyallocation plans are considered: i) 5 MHz spectrum allocated only to theLP stations; ii) 9 MHz to HP transmitters and 1 MHz to LP transmitters.Note that lower bandwidth is typically allocated to LP stations due totheir DSL backhaul connectivity with limited throughput. FIGS. 22a-bshows that when using 50% of HP transmitters it is possible to increasesignificantly the rate distribution, raising the average per-client datarate from 2.4 Mbps in FIGS. 22a to 38 Mbps in FIG. 22 b.

Next, we defined algorithms to control power transmission of LP stationssuch that higher power is allowed at any given time, thereby increasingthe throughput over the downlink channel of DIDO systems in FIG. 22b .We observe that the FCC limits on the power density is defined based onaverage over time as [2]

$\begin{matrix}{S = \frac{\sum_{n = 1}^{N}{S_{n}t_{n}}}{T_{MPE}}} & (14)\end{matrix}$where T_(MPE)=Σ_(n=1) ^(N) t_(n) is the MPE averaging time, t_(n) is theperiod of time of exposure to radiation with power density S_(n). For“controlled” exposure the average time is 6 minutes, whereas for“uncontrolled” exposure it is increased up to 30 minutes. Then, anypower source is allowed to transmit at larger power levels than the MPElimits, as long as the average power density in (14) satisfies the FCClimit over 30 minute average for “uncontrolled” exposure.

Based on this analysis, we define adaptive power control methods toincrease instantaneous per-antenna transmit power, while maintainingaverage power per DIDO antenna below MPE limits. We consider DIDOsystems with more transmit antennas than active clients. This is areasonable assumption given that DIDO antennas can be conceived asinexpensive wireless devices (similar to WiFi access points) and can beplaced anywhere there is DSL, cable modem, optical fiber, or otherInternet connectivity.

The framework of DIDO systems with adaptive per-antenna power control isdepicted in FIG. 23. The amplitude of the digital signal coming out ofthe multiplexer 234 is dynamically adjusted with power scaling factorsS₁, . . . , S_(M), before being sent to the DAC units 235. The powerscaling factors are computed by the power control unit 232 based on theCQI 233.

In one embodiment, N_(g) DIDO antenna groups are defined. Every groupcontains at least as many DIDO antennas as the number of active clients(K). At any given time, only one group has N_(a)>K active DIDO antennastransmitting to the clients at larger power level (S_(o)) than MPE limit(MPE). One method iterates across all antenna groups according toRound-Robin scheduling policy depicted in FIG. 24. In anotherembodiment, different scheduling techniques (i.e., proportional-fairscheduling [8]) are employed for cluster selection to optimize errorrate or throughput performance.

Assuming Round-Robin power allocation, from (14) we derive the averagetransmit power for every DIDO antenna as

$\begin{matrix}{S = {{S_{o}\frac{t_{o}}{T_{MPE}}} \leq \overset{\_}{MPE}}} & (15)\end{matrix}$where t_(o) is the period of time over which the antenna group is activeand T_(MPE)=30 min is the average time defined by the FCC guidelines[2]. The ratio in (15) is the duty factor (DF) of the groups, definedsuch that the average transmit power from every DIDO antenna satisfiesthe MPE limit (MPE). The duty factor depends on the number of activeclients, the number of groups and active antennas per-group, accordingto the following definition

$\begin{matrix}{{DF}\overset{\Delta}{=}{\frac{K}{N_{g}N_{a}} = {\frac{t_{o}}{T_{{MPE}\;}}.}}} & (16)\end{matrix}$The SNR gain (in dB) obtained in DIDO systems with power control andantenna grouping is expressed as a function of the duty factor as

$\begin{matrix}{G_{d\; B} = {10{{\log_{10}( \frac{1}{DF} )}.}}} & (17)\end{matrix}$We observe the gain in (17) is achieved at the expense of G_(dB)additional transmit power across all DIDO antennas.In general, the total transmit power from all N_(a) of all N_(g) groupsis defined asP=Σ _(j=1) ^(N) ^(g) Σ_(i=1) ^(N) ^(a) P _(ij)  (18)where the P_(ij) is the average per-antenna transmit power given by

$\begin{matrix}{P_{ij} = {{\frac{1}{T_{MPE}}{\int_{0}^{T_{MPE}}{{S_{ij}(t)}d\; t}}} \leq \overset{\_}{MPE}}} & (19)\end{matrix}$and S_(ij)(t) is the power spectral density for the i^(th) transmitantenna within the j^(th) group. In one embodiment, the power spectraldensity in (19) is designed for every antenna to optimize error rate orthroughput performance.

To gain some intuition on the performance of the proposed method,consider 400 DIDO distributed antennas in a given coverage area and 400clients subscribing to a wireless Internet service offered over DIDOsystems. It is unlikely that every Internet connection will be fullyutilized all the time. Let us assume that 10% of the clients will beactively using the wireless Internet connection at any given time. Then,400 DIDO antennas can be divided in N_(g)=10 groups of N_(a)=40 antennaseach, every group serving K=40 active clients at any given time withduty factor DF=0.1. The SNR gain resulting from this transmission schemeis G_(dB)=10 log₁₀(1/DF)=10 dB, provided by 10 dB additional transmitpower from all DIDO antennas. We observe, however, that the averageper-antenna transmit power is constant and is within the MPE limit.

FIG. 25 compares the (uncoded) SER performance of the above powercontrol with antenna grouping against conventional eigenmode selectionin U.S. Pat. No. 7,636,381. All schemes use BD precoding with fourclients, each client equipped with single antenna. The SNR refers to theratio of per-transmit-antenna power over noise power (i.e., per-antennatransmit SNR). The curve denoted with DIDO 4×4 assumes four transmitantenna and BD precoding. The curve with squares denotes the SERperformance with two extra transmit antennas and BD with eigenmodeselection, yielding 10 dB SNR gain (at 1% SER target) over conventionalBD precoding. Power control with antenna grouping and DF=1/10 yields 10dB gain at the same SER target as well. We observe that eigenmodeselection changes the slope of the SER curve due to diversity gain,whereas our power control method shifts the SER curve to the left(maintaining the same slope) due to increased average transmit power.For comparison, the SER with larger duty factor DF=1/50 is shown toprovide additional 7 dB gain compared to DF=1/10.

Note that our power control may have lower complexity than conventionaleigenmode selection methods. In fact, the antenna ID of every group canbe pre-computed and shared among DIDO antennas and clients via lookuptables, such that only K channel estimates are required at any giventime. For eigenmode selection, (K+2) channel estimates are computed andadditional computational processing is required to select the eigenmodethat minimizes the SER at any given time for all clients.

Next, we describe another method involving DIDO antenna grouping toreduce CSI feedback overhead in some special scenarios. FIG. 26a showsone scenario where clients (dots) are spread randomly in one areacovered by multiple DIDO distributed antennas (crosses). The averagepower over every transmit-receive wireless link can be computed asA={|H| ²}.  (20)where H is the channel estimation matrix available at the DIDO BTS.

The matrices A in FIGS. 26a-c are obtained numerically by averaging thechannel matrices over 1000 instances. Two alternative scenarios aredepicted in FIG. 26b and FIG. 26c , respectively, where clients aregrouped together around a subset of DIDO antennas and receive negligiblepower from DIDO antennas located far away. For example, FIG. 26b showstwo groups of antennas yielding block diagonal matrix A. One extremescenario is when every client is very close to only one transmitter andthe transmitters are far away from one another, such that the power fromall other DIDO antennas is negligible. In this case, the DIDO linkdegenerates in multiple SISO links and A is a diagonal matrix as in FIG.26 c.

In all three scenarios above, the BD precoding dynamically adjusts theprecoding weights to account for different power levels over thewireless links between DIDO antennas and clients. It is convenient,however, to identify multiple groups within the DIDO cluster and operateDIDO precoding only within each group. Our proposed grouping methodyields the following advantages:

-   -   Computational gain: DIDO precoding is computed only within every        group in the cluster. For example, if BD precoding is used,        singular value decomposition (SVD) has complexity O(n³), where n        is the minimum dimension of the channel matrix H. If H can be        reduced to a block diagonal matrix, the SVD is computed for        every block with reduced complexity. In fact, if the channel        matrix is divided into two block matrices with dimensions n₁ and        n₂ such that n=n₁+n₂, the complexity of the SVD is only O(n₁        ³)+O(n₂ ³)<O(n³). In the extreme case, if H is diagonal matrix,        the DIDO link reduce to multiple SISO links and no SVD        calculation is required.    -   Reduced CSI feedback overhead: When DIDO antennas and clients        are divided into groups, in one embodiment, the CSI is computed        from the clients to the antennas only within the same group. In        TDD systems, assuming channel reciprocity, antenna grouping        reduces the number of channel estimates to compute the channel        matrix H. In FDD systems where the CSI is fed back over the        wireless link, antenna grouping further yields reduction of CSI        feedback overhead over the wireless links between DIDO antennas        and clients.        Multiple Access Techniques for the DIDO Uplink Channel

In one embodiment of the invention, different multiple access techniquesare defined for the DIDO uplink channel. These techniques can be used tofeedback the CSI or transmit data streams from the clients to the DIDOantennas over the uplink. Hereafter, we refer to feedback CSI and datastreams as uplink streams.

-   -   Multiple-input multiple-output (MIMO): the uplink streams are        transmitted from the client to the DIDO antennas via open-loop        MIMO multiplexing schemes. This method assumes all clients are        time/frequency synchronized. In one embodiment, synchronization        among clients is achieved via training from the downlink and all        DIDO antennas are assumed to be locked to the same        time/frequency reference clock. Note that variations in delay        spread at different clients may generate jitter between the        clocks of different clients that may affect the performance of        MIMO uplink scheme. After the clients send uplink streams via        MIMO multiplexing schemes, the receive DIDO antennas may use        non-linear (i.e., maximum likelihood, ML) or linear (i.e.,        zeros-forcing, minimum mean squared error) receivers to cancel        co-channel interference and demodulate the uplink streams        individually.    -   Time division multiple access (TDMA): Different clients are        assigned to different time slots. Every client sends its uplink        stream when its time slot is available.    -   Frequency division multiple access (FDMA): Different clients are        assigned to different carrier frequencies. In multicarrier        (OFDM) systems, subsets of tones are assigned to different        clients that transmit the uplink streams simultaneously, thereby        reducing latency.    -   Code division multiple access (CDMA): Every client is assigned        to a different pseudo-random sequence and orthogonality across        clients is achieved in the code domain.

In one embodiment of the invention, the clients are wireless devicesthat transmit at much lower power than the DIDO antennas. In this case,the DIDO BTS defines client sub-groups based on the uplink SNRinformation, such that interference across sub-groups is minimized.Within every sub-group, the above multiple access techniques areemployed to create orthogonal channels in time, frequency, space or codedomains thereby avoiding uplink interference across different clients.

In another embodiment, the uplink multiple access techniques describedabove are used in combination with antenna grouping methods presented inthe previous section to define different client groups within the DIDOcluster.

System and Method for Link Adaptation in DIDO Multicarrier Systems

Link adaptation methods for DIDO systems exploiting time, frequency andspace selectivity of wireless channels were defined in U.S. Pat. No.7,636,381. Described below are embodiments of the invention for linkadaptation in multicarrier (OFDM) DIDO systems that exploittime/frequency selectivity of wireless channels.

We simulate Rayleigh fading channels according to the exponentiallydecaying power delay profile (PDP) or Saleh-Valenzuela model in [9]. Forsimplicity, we assume single-cluster channel with multipath PDP definedasP _(n) =e ^(−βn)  (21)where n=0, . . . , L−1, is the index of the channel tap, L is the numberof channel taps and β=1/σ_(DS) is the PDP exponent that is an indicatorof the channel coherence bandwidth, inverse proportional to the channeldelay spread (σ_(DS)). Low values of β yield frequency-flat channels,whereas high values of β produce frequency selective channels. The PDPin (21) is normalized such that the total average power for all Lchannel taps is unitary

$\begin{matrix}{\overset{\_}{P_{n}} = {\frac{P_{n}}{\sum_{i = 0}^{L - 1}P_{i}}.}} & (22)\end{matrix}$FIG. 27 depicts the amplitude of low frequency selective channels(assuming β=1) over delay domain or instantaneous PDP (upper plot) andfrequency domain (lower plot) for DIDO 2×2 systems. The first subscriptindicates the client, the second subscript the transmit antenna. Highfrequency selective channels (with β=0.1) are shown in FIG. 28.

Next, we study the performance of DIDO precoding in frequency selectivechannels. We compute the DIDO precoding weights via BD, assuming thesignal model in (1) that satisfies the condition in (2). We reformulatethe DIDO receive signal model in (5), with the condition in (2), asr _(k) =H _(ek) S _(k) +n _(k).  (23)

where H_(ex)=H_(k)W_(k) is the effective channel matrix for user k. ForDIDO 2×2, with a single antenna per client, the effective channel matrixreduces to one value with a frequency response shown in FIG. 29 and forchannels characterized by high frequency selectivity (e.g., with β=0.1)in FIG. 28. The continuous line in FIG. 29 refers to client 1, whereasthe line with dots refers to client 2. Based on the channel qualitymetric in FIG. 29 we define time/frequency domain link adaptation (LA)methods that dynamically adjust MCSs, depending on the changing channelconditions.

We begin by evaluating the performance of different MCSs in AWGN andRayleigh fading SISO channels. For simplicity, we assume no FEC coding,but the following LA methods can be extended to systems that includeFEC.

FIG. 30 shows the SER for different QAM schemes (i.e., 4-QAM, 16-QAM,64-QAM). Without loss of generality, we assume target SER of 1% foruncoded systems. The SNR thresholds to meet that target SER in AWGNchannels are 8 dB, 15.5 dB and 22 dB for the three modulation schemes,respectively. In Rayleigh fading channels, it is well known the SERperformance of the above modulation schemes is worse than AWGN [13] andthe SNR thresholds are: 18.6 dB, 27.3 dB and 34.1 dB, respectively. Weobserve that DIDO precoding transforms the multi-user downlink channelinto a set of parallel SISO links. Hence, the same SNR thresholds as inFIG. 30 for SISO systems hold for DIDO systems on a client-by-clientbasis. Moreover, if instantaneous LA is carried out, the thresholds inAWGN channels are used.

The key idea of the proposed LA method for DIDO systems is to use lowMCS orders when the channel undergoes deep fades in the time domain orfrequency domain (depicted in FIG. 28) to provide link-robustness.Contrarily, when the channel is characterized by large gain, the LAmethod switches to higher MCS orders to increase spectral efficiency.One contribution of the present application compared to U.S. Pat. No.7,636,381 is to use the effective channel matrix in (23) and in FIG. 29as a metric to enable adaptation.

The general framework of the LA methods is depicted in FIG. 31 anddefined as follows:

-   -   CSI estimation: At 3171 the DIDO BTS computes the CSI from all        users. Users may be equipped with single or multiple receive        antennas.    -   DIDO precoding: At 3172, the BTS computes the DIDO precoding        weights for all users. In one embodiment, BD is used to compute        these weights. The precoding weights are calculated on a        tone-by-tone basis.    -   Link-quality metric calculation: At 3173 the BTS computes the        frequency-domain link quality metrics. In OFDM systems, the        metrics are calculated from the CSI and DIDO precoding weights        for every tone. In one embodiment of the invention, the        link-quality metric is the average SNR over all OFDM tones. We        define this method as LA1 (based on average SNR performance). In        another embodiment, the link quality metric is the frequency        response of the effective channel in (23). We define this method        as LA2 (based on tone-by-tone performance to exploit frequency        diversity). If every client has single antenna, the        frequency-domain effective channel is depicted in FIG. 29. If        the clients have multiple receive antennas, the link-quality        metric is defined as the Frobenius norm of the effective channel        matrix for every tone. Alternatively, multiple link-quality        metrics are defined for every client as the singular values of        the effective channel matrix in (23).    -   Bit-loading algorithm: At 3174, based on the link-quality        metrics, the BTS determines the MCSs for different clients and        different OFDM tones. For LA1 method, the same MCS is used for        all clients and all OFDM tones based on the SNR thresholds for        Rayleigh fading channels in FIG. 30. For LA2, different MCSs are        assigned to different OFDM tones to exploit channel frequency        diversity.    -   Precoded data transmission: At 3175, the BTS transmits precoded        data streams from the DIDO distributed antennas to the clients        using the MCSs derived from the bit-loading algorithm. One        header is attached to the precoded data to communicate the MCSs        for different tones to the clients. For example, if eight MCSs        are available and the OFDM symbols are defined with N=64 tone,        log₂(8)*N=192 bits are required to communicate the current MCS        to every client. Assuming 4-QAM (2 bits/symbol spectral        efficiency) is used to map those bits into symbols, only        192/2/N=1.5 OFDM symbols are required to map the MCS        information. In another embodiment, multiple subcarriers (or        OFDM tones) are grouped into subbands and the same MCS is        assigned to all tones in the same subband to reduce the overhead        due to control information. Moreover, the MCS are adjusted based        on temporal variations of the channel gain (proportional to the        coherence time). In fixed-wireless channel (characterized by low        Doppler effect) the MCS are recalculated every fraction of the        channel coherence time, thereby reducing the overhead required        for control information.

FIG. 32 shows the SER performance of the LA methods described above. Forcomparison, the SER performance in Rayleigh fading channels is plottedfor each of the three QAM schemes used. The LA2 method adapts the MCSsto the fluctuation of the effective channel in the frequency domain,thereby providing 1.8 bps/Hz gain in spectral efficiency for low SNR(i.e., SNR=20 dB) and 15 dB gain in SNR (for SNR>35 dB) compared to LA1.

System and Method for DIDO Precoding Interpolation in MulticarrierSystems

The computational complexity of DIDO systems is mostly localized at thecentralized processor or BTS. The most computationally expensiveoperation is the calculation of the precoding weights for all clientsfrom their CSI. When BD precoding is employed, the BTS has to carry outas many singular value decomposition (SVD) operations as the number ofclients in the system. One way to reduce complexity is throughparallelized processing, where the SVD is computed on a separateprocessor for every client.

In multicarrier DIDO systems, each subcarrier undergoes flat-fadingchannel and the SVD is carried out for every client over everysubcarrier. Clearly the complexity of the system increases linearly withthe number of subcarriers. For example, in OFDM systems with 1 MHzsignal bandwidth, the cyclic prefix (L₀) must have at least eightchannel taps (i.e., duration of 8 microseconds) to avoid intersymbolinterference in outdoor urban macrocell environments with large delayspread [3]. The size (N_(FFT)) of the fast Fourier transform (FFT) usedto generate the OFDM symbols is typically set to multiple of L₀ toreduce loss of data rate. If N_(FFT)=64, the effective spectralefficiency of the system is limited by a factorN_(FFT)/(N_(FFT)+L₀)=89%. Larger values of N_(FFT) yield higher spectralefficiency at the expense of higher computational complexity at the DIDOprecoder.

One way to reduce computational complexity at the DIDO precoder is tocarry out the SVD operation over a subset of tones (that we call pilottones) and derive the precoding weights for the remaining tones viainterpolation. Weight interpolation is one source of error that resultsin inter-client interference. In one embodiment, optimal weightinterpolation techniques are employed to reduce inter-clientinterference, yielding improved error rate performance and lowercomputational complexity in multicarrier systems. In DIDO systems with Mtransmit antennas, U clients and N receive antennas per clients, thecondition for the precoding weights of the k^(th) client (W_(k)) thatguarantees zero interference to the other clients u is derived from (2)asH _(u) w _(k)=0^(N×N) ; ∀u=1, . . . ,U; with u≠k  (24)where H_(u) are the channel matrices corresponding to the other DIDOclients in the system.

In one embodiment of the invention, the objective function of the weightinterpolation method is defined as

$\begin{matrix}{{f( \theta_{k} )} = {\sum_{\underset{u \neq k}{u = 1}}^{U}{{H_{u}{{\hat{W}}_{k}( \theta_{k} )}}}_{F}}} & (25)\end{matrix}$where θ_(k) is the set of parameters to be optimized for user k,Ŵ_(k)(θ_(k)) is the weight interpolation matrix and ∥·∥_(F) denotes theFrobenius norm of a matrix. The optimization problem is formulated as

$\begin{matrix}{\theta_{k,{opt}} = {\arg\;{\min\limits_{\theta_{k}{\epsilon\Theta}_{k}}{f( \theta_{k} )}}}} & (26)\end{matrix}$where Θ_(k) is the feasible set of the optimization problem andθ_(k,opt) is the optimal solution.

The objective function in (25) is defined for one OFDM tone. In anotherembodiment of the invention, the objective function is defined as linearcombination of the Frobenius norm in (25) of the matrices for all theOFDM tones to be interpolated. In another embodiment, the OFDM spectrumis divided into subsets of tones and the optimal solution is given by

$\begin{matrix}{\theta_{k,{opt}} = {\arg\;{\min\limits_{\theta_{k}{\epsilon\Theta}_{k}}{\max\limits_{n\;\epsilon\; A}{f( {n,\theta_{k}} )}}}}} & (27)\end{matrix}$where n is the OFDM tone index and A is the subset of tones.

The weight interpolation matrix W_(k)(θ_(k)) in (25) is expressed as afunction of a set of parameters θ_(k). Once the optimal set isdetermined according to (26) or (27), the optimal weight matrix iscomputed. In one embodiment of the invention, the weight interpolationmatrix of given OFDM tone n is defined as linear combination of theweight matrices of the pilot tones. One example of weight interpolationfunction for beamforming systems with single client was defined in [11].In DIDO multi-client systems we write the weight interpolation matrix asŴ _(k)(lN ₀ +n,θ _(k))=(1−c _(n))·W(l)+c _(n) e ^(jθ) ^(k) ·W(l+1)  (28)where 0≤l≤(L₀−1), L₀ is the number of pilot tones and c_(n)=(n−1)/N₀,with N₀=N_(FFT)/L₀. The weight matrix in (28) is then normalized suchthat ∥Ŵ∥_(F)=√{square root over (NM)} to guarantee unitary powertransmission from every antenna. If N=1 (single receive antenna perclient), the matrix in (28) becomes a vector that is normalized withrespect to its norm. In one embodiment of the invention, the pilot tonesare chosen uniformly within the range of the OFDM tones. In anotherembodiment, the pilot tones are adaptively chosen based on the CSI tominimize the interpolation error.

We observe that one key difference of the system and method in [11]against the one proposed in this patent application is the objectivefunction. In particular, the systems in [11] assumes multiple transmitantennas and single client, so the related method is designed tomaximize the product of the precoding weight by the channel to maximizethe receive SNR for the client. This method, however, does not work inmulti-client scenarios, since it yields inter-client interference due tointerpolation error. By contrast, our method is designed to minimizeinter-client interference thereby improving error rate performance toall clients.

FIG. 33 shows the entries of the matrix in (28) as a function of theOFDM tone index for DIDO 2×2 systems with N_(FFT)=64 and L₀=8. Thechannel PDP is generated according to the model in (21) with β=1 and thechannel consists of only eight channel taps. We observe that L₀ must bechosen to be larger than the number of channel taps. The solid lines inFIG. 33 represent the ideal functions, whereas the dotted lines are theinterpolated ones. The interpolated weights match the ideal ones for thepilot tones, according to the definition in (28). The weights computedover the remaining tones only approximate the ideal case due toestimation error.

One way to implement the weight interpolation method is via exhaustivesearch over the feasible set Θ_(k) in (26). To reduce the complexity ofthe search, we quantize the feasible set into P values uniformly in therange [0,2π]. FIG. 34 shows the SER versus SNR for L₀=8, M=N_(t)=2transmit antennas and variable number of P. As the number ofquantization levels increases, the SER performance improves. We observethe case P=10 approaches the performance of P=100 for much lowercomputational complexity, due to reduced number of searches.

FIG. 35 shows the SER performance of the interpolation method fordifferent DIDO orders and L₀=16. We assume the number of clients is thesame as the number of transmit antennas and every client is equippedwith single antenna. As the number of clients increases the SERperformance degrades due to increase inter-client interference producedby weight interpolation errors.

In another embodiment of the invention, weight interpolation functionsother than those in (28) are used. For example, linear predictionautoregressive models [12] can be used to interpolate the weights acrossdifferent OFDM tones, based on estimates of the channel frequencycorrelation.

REFERENCES

-   [1] A. Forenza and S. G. Perlman, “System and method for distributed    antenna wireless communications”, U.S. application Ser. No.    12/630,627, filed Dec. 2, 2009, entitled “System and Method For    Distributed Antenna Wireless Communications”-   [2] FCC, “Evaluating compliance with FCC guidelines for human    exposure to radiofrequency electromagnetic fields,” OET Bulletin 65,    Ed. 97-01, August 1997-   [3] 3GPP, “Spatial Channel Model AHG (Combined ad-hoc from 3GPP &    3GPP2)”, SCM Text V6.0, Apr. 22, 2003-   [4] 3GPP TR 25.912, “Feasibility Study for Evolved UTRA and UTRAN”,    V9.0.0 (2009-10)-   [5] 3GPP TR 25.913, “Requirements for Evolved UTRA (E-UTRA) and    Evolved UTRAN (E-UTRAN)”, V8.0.0 (2009-01)-   [6] W. C. Jakes, Microwave Mobile Communications, IEEE Press, 1974-   [7] K. K. Wong, et al., “A joint channel diagonalization for    multiuser MIMO antenna systems,” IEEE Trans. Wireless Comm., vol. 2,    pp. 773-786, July 2003;-   [8] P. Viswanath, et al., “Opportunistic beamforming using dump    antennas,” IEEE Trans. On Inform. Theory, vol. 48, pp. 1277-1294,    June 2002.-   [9] A. A. M. Saleh, et al., “A statistical model for indoor    multipath propagation,” IEEE Jour. Select. Areas in Comm., vol. 195    SAC-5, no. 2, pp. 128-137, February 1987.-   [10] A. Paulraj, et al., Introduction to Space-Time Wireless    Communications, Cambridge University Press, 40 West 20th Street, New    York, N.Y., USA, 2003.-   [11] J. Choi, et al., “Interpolation Based Transmit Beamforming for    MIMO-OFDM with Limited Feedback,” IEEE Trans. on Signal Processing,    vol. 53, no. 11, pp. 4125-4135, November 2005.-   [12] I. Wong, et al., “Long Range Channel Prediction for Adaptive    OFDM Systems,” Proc. of the IEEE Asilomar Conf. on Signals, Systems,    and Computers, vol. 1, pp. 723-736, Pacific Grove, Calif., USA, Nov.    7-10, 2004.-   [13] J. G. Proakis, Communication System Engineering, Prentice Hall,    1994-   [14] B. D. Van Veen, et al., “Beamforming: a versatile approach to    spatial filtering,” IEEE ASSP Magazine, April 1988.-   [15] R. G. Vaughan, “On optimum combining at the mobile,” IEEE    Trans. On Vehic. Tech., vol 37, n. 4, pp. 181-188, November 1988-   [16] F. Qian, “Partially adaptive beamforming for correlated    interference rejection,” IEEE Trans. On Sign. Proc., vol. 43, n. 2,    pp. 506-515, February 1995-   [17] H. Krim, et. al., “Two decades of array signal processing    research,” IEEE Signal Proc. Magazine, pp. 67-94, July 1996-   [19] W. R. Remley, “Digital beamforming system”, U.S. Pat. No.    4,003,016, January 1977-   [18] R. J. Masak, “Beamforming/null-steering adaptive array”, U.S.    Pat. No. 4,771,289, September 1988-   [20] K.-B. Yu, et. al., “Adaptive digital beamforming architecture    and algorithm for nulling mainlobe and multiple sidelobe radar    jammers while preserving monopulse ratio angle estimation accuracy”,    U.S. Pat. No. 5,600,326, February 1997-   [21] H. Boche, et al., “Analysis of different precoding/decoding    strategies for multiuser beamforming”, IEEE Vehic. Tech. Conf., vol.    1, April 2003-   [22] M. Schubert, et al., “Joint ‘dirty paper’ pre-coding and    downlink beamforming,” vol. 2, pp. 536-540, December 2002-   [23] H. Boche, et al. “A general duality theory for uplink and    downlink beamformingc”, vol. 1, pp. 87-91, December 2002-   [24] K. K. Wong, R. D. Murch, and K. B. Letaief, “A joint channel    diagonalization for multiuser MIMO antenna systems,” IEEE Trans.    Wireless Comm., vol. 2, pp. 773-786, July 2003;-   [25] Q. H. Spencer, A. L. Swindlehurst, and M. Haardt, “Zero forcing    methods for downlink spatial multiplexing in multiuser MIMO    channels,” IEEE Trans. Sig. Proc., vol. 52, pp. 461-471, February    2004.    II. Disclosure from Related Application Ser. No. 12/917,257

Described below are wireless radio frequency (RF) communication systemsand methods employing a plurality of distributed transmitting antennasoperating cooperatively to create wireless links to given users, whilesuppressing interference to other users. Coordination across differenttransmitting antennas is enabled via user-clustering. The user clusteris a subset of transmitting antennas whose signal can be reliablydetected by given user (i.e., received signal strength above noise orinterference level). Every user in the system defines its ownuser-cluter. The waveforms sent by the transmitting antennas within thesame user-cluster coherently combine to create RF energy at the targetuser's location and points of zero RF interference at the location ofany other user reachable by those antennas.

Consider a system with M transmit antennas within one user-cluster and Kusers reachable by those M antennas, with K≤M. We assume thetransmitters are aware of the CSI (H∈C^(K×M)) between the M transmitantennas and K users. For simplicity, every user is assumed to beequipped with a single antenna, but the same method can be extended tomultiple receive antennas per user. Consider the channel matrix Hobtained by combining the channel vectors (h_(k)∈C^(1×M)) from the Mtransmit antennas to the K users as

$H = {\begin{bmatrix}h_{1} \\\vdots \\h_{k} \\\vdots \\h_{K}\end{bmatrix}.}$The precoding weights (w_(k)∈C^(M×1)) that create RF energy to user kand zero RF energy to all other K−1 users are computed to satisfy thefollowing condition{tilde over (H)} _(k) w _(k)=0^(K×1)where {tilde over (H)}_(k) is the effective channel matrix of user kobtained by removing the k-th row of matrix H and 0^(K×1) is the vectorwith all zero entries

In one embodiment, the wireless system is a DIDO system and userclustering is employed to create a wireless communication link to thetarget user, while pre-cancelling interference to any other userreachable by the antennas lying within the user-cluster. In U.S.application Ser. No. 12/630,627, a DIDO system is described whichincludes:

-   -   DIDO clients: user terminals equipped with one or multiple        antennas;    -   DIDO distributed antennas: transceiver stations operating        cooperatively to transmit precoded data streams to multiple        users, thereby suppressing inter-user interference;    -   DIDO base transceiver stations (BTS): centralized processor        generating precoded waveforms to the DIDO distributed antennas;    -   DIDO base station network (BSN): wired backhaul connecting the        BTS to the DIDO distributed antennas or to other BTSs.        -   The DIDO distributed antennas are grouped into different            subsets depending on their spatial distribution relative to            the location of the BTSs or DIDO clients. We define three            types of clusters, as depicted in FIG. 36:    -   Super-cluster 3640: is the set of DIDO distributed antennas        connected to one or multiple BTSs such that the round-trip        latency between all BTSs and the respective users is within the        constraint of the DIDO precoding loop;    -   DIDO-cluster 3641: is the set of DIDO distributed antennas        connected to the same BTS. When the super-cluster contains only        one BTS, its definition coincides with the DIDO-cluster;    -   User-cluster 3642: is the set of DIDO distributed antennas that        cooperatively transmit precoded data to given user.

For example, the BTSs are local hubs connected to other BTSs and to theDIDO distributed antennas via the BSN. The BSN can be comprised ofvarious network technologies including, but not limited to, digitalsubscriber lines (DSL), ADSL, VDSL [6], cable modems, fiber rings, T1lines, hybrid fiber coaxial (HFC) networks, and/or fixed wireless (e.g.,WiFi). All BTSs within the same super-cluster share information aboutDIDO precoding via the BSN such that the round-trip latency is withinthe DIDO precoding loop.

In FIG. 37, the dots denote DIDO distributed antennas, the crosses arethe users and the dashed lines indicate the user-clusters for users U1and U8, respectively. The method described hereafter is designed tocreate a communication link to the target user U1 while creating pointsof zero RF energy to any other user (U2-U8) inside or outside theuser-cluster.

We proposed similar method in [5], where points of zero RF energy werecreated to remove interference in the overlapping regions between DIDOclusters. Extra antennas were required to transmit signal to the clientswithin the DIDO cluster while suppressing inter-cluster interference.One embodiment of a method proposed in the present application does notattempt to remove inter-DIDO-cluster interference; rather it assumes thecluster is bound to the client (i.e., user-cluster) and guarantees thatno interference (or negligible interference) is generated to any otherclient in that neighborhood.

One idea associated with the proposed method is that users far enoughfrom the user-cluster are not affected by radiation from the transmitantennas, due to large pathloss. Users close or within the user-clusterreceive interference-free signal due to precoding. Moreover, additionaltransmit antennas can be added to the user-cluster (as shown in FIG. 37)such that the condition K≤M is satisfied.

One embodiment of a method employing user clustering consists of thefollowing steps:

a. Link-quality measurements: the link quality between every DIDOdistributed antenna and every user is reported to the BTS. Thelink-quality metric consists of signal-to-noise ratio (SNR) orsignal-to-interference-plus-noise ratio (SINR). In one embodiment, theDIDO distributed antennas transmit training signals and the usersestimate the received signal quality based on that training. Thetraining signals are designed to be orthogonal in time, frequency orcode domains such that the users can distinguish across differenttransmitters. Alternatively, the DIDO antennas transmit narrowbandsignals (i.e., single tone) at one particular frequency (i.e., a beaconchannel) and the users estimate the link-quality based on that beaconsignal. One threshold is defined as the minimum signal amplitude (orpower) above the noise level to demodulate data successfully as shown inFIG. 38a . Any link-quality metric value below this threshold is assumedto be zero. The link-quality metric is quantized over a finite number ofbits and fed back to the transmitter.In a different embodiment, the training signals or beacons are sent fromthe users and the link quality is estimated at the DIDO transmitantennas (as in FIG. 38b ), assuming reciprocity between uplink (UL) anddownlink (DL) pathloss. Note that pathloss reciprocity is a realisticassumption in time division duplexing (TDD) systems (with UL and DLchannels at the same frequency) and frequency division duplexing (FDD)systems when the UL and DL frequency bands are reatively close.Information about the link-quality metrics is shared across differentBTSs through the BSN as depicted in FIG. 37 such that all BTSs are awareof the link-quality between every antenna/user couple across differentDIDO clusters.b. Definition of user-clusters: the link-quality metrics of all wirelesslinks in the DIDO clusters are the entries to the link-quality matrixshared across all BTSs via the BSN. One example of link-quality matrixfor the scenario in FIG. 37 is depicted in FIG. 39.The link-quality matrix is used to define the user clusters. Forexample, FIG. 39 shows the selection of the user cluster for user U8.The subset of transmitters with non-zero link-quality metrics (i.e.,active transmitters) to user U8 is first identified. These transmitterspopulate the user-cluster for the user U8. Then the sub-matrixcontaining non-zero entries from the transmitters within theuser-cluster to the other users is selected. Note that since thelink-quality metrics are only used to select the user cluster, they canbe quantized with only two bits (i.e., to identify the state above orbelow the thresholds in FIG. 38) thereby reducing feedback overhead.

Another example is depicted in FIG. 40 for user U1. In this case thenumber of active transmitters is lower than the number of users in thesub-matrix, thereby violating the condition K≤M. Therefore, one or morecolumns are added to the sub-matrix to satisfy that condition. If thenumber of transmitters exceeds the number of users, the extra antennascan be used for diversity schemes (i.e., antenna or eigenmodeselection).

Yet another example is shown in FIG. 41 for user U4. We observe that thesub-matrix can be obtained as combination of two sub-matrices.

c. CSI report to the BTSs: Once the user clusters are selected, the CSIfrom all transmitters within the user-cluster to every user reached bythose transmitters is made available to all BTSs. The CSI information isshared across all BTSs via the BSN. In TDD systems, UL/DL channelreciprocity can be exploited to derive the CSI from training over the ULchannel. In FDD systems, feedback channels from all users to the BTSsare required. To reduce the amount of feedback, only the CSIcorresponding to the non-zero entries of the link-quality matrix are fedback.d. DIDO precoding: Finally, DIDO precoding is applied to every CSIsub-matrix corresponding to different user clusters (as described, forexample, in the related U.S. patent applications).In one embodiment, singular value decomposition (SVD) of the effectivechannel matrix {tilde over (H)}_(k) is computed and the precoding weightw_(k) for user k is defined as the right singular vector correspondingto the null subspace of {tilde over (H)}_(k). Alternatively, if M>K andthe SVD decomposes the effective channel matrix as {tilde over(H)}_(k)=V_(k)Σ_(k)U_(k) ^(H), the DIDO precoding weight for user k isgiven byW _(k) =U _(o)(U _(o) ^(H) ·h _(k) ^(T))where U_(o) is the matrix with columns being the singular vectors of thenull subspace of {tilde over (H)}_(k).From basic linear algebra considerations, we observe that the rightsingular vector in the null subspace of the matrix {tilde over (H)} isequal to the eigenvetor of C corresponding to the zero eigenvalueC={tilde over (H)} ^(H) {tilde over (H)}=(VΣU ^(H))^(H)(VΣU ^(H))=UΣ ² U^(H)where the effective channel matrix is decomposed as {tilde over(H)}=VΣU^(H), according to the SVD. Then, one alternative to computingthe SVD of {tilde over (H)}_(K) is to calculate the eigenvaluedecomposition of C. There are several methods to compute eigenvaluedecomposition such as the power method. Since we are only interested tothe eigenvector corresponding to the null subspace of C, we use theinverse power method described by the iteration

$u_{i + 1} = \frac{( {C - {\lambda\; I}} )^{- 1}u_{i}}{{( {C - {\lambda\; I}} )^{- 1}u_{i}}}$where the vector (u_(i)) at the first iteration is a random vector.

Given that the eigenvalue (λ) of the null subspace is known (i.e., zero)the inverse power method requires only one iteration to converge,thereby reducing computational complexity. Then, we write the precodingweight vector asw=C ⁻¹ u ₁where u₁ is the vector with real entries equal to 1 (i.e., the precodingweight vector is the sum of the columns of c⁻¹).The DIDO precoding calculation requires one matrix inversion. There areseveral numerical solutions to reduce the complexity of matrixinversions such as the Strassen's algorithm [1] or theCoppersmith-Winograd's algorithm [2,3]. Since Cis Hermitian matrix bydefinition, an alternative solution is to decompose C in its real andimaginary components and compute matrix inversion of a real matrix,according to the method in [4. Section 11.4].

Another feature of the proposed method and system is itsreconfigurability. As the client moves across different DIDO clusters asin FIG. 42, the user-cluster follows its moves. In other words, thesubset of transmit antennas is constantly updated as the client changesits position and the effective channel matrix (and correspondingprecoding weights) are recomputed.

The method proposed herein works within the super-cluster in FIG. 36,since the links between the BTSs via the BSN must be low-latency. Tosuppress interference in the overlapping regions of differentsuper-clusters, it is possible to use our method in [5] that uses extraantennas to create points of zero RF energy in the interfering regionsbetween DIDO clusters.

It should be noted that the terms “user” and “client” are usedinterchangeably herein.

REFERENCES

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The capacity of multiple antenna systems (MAS) in practical propagationenvironments is a function of the spatial diversity available over thewireless link. Spatial diversity is determined by the distribution ofscattering objects in the wireless channel as well as the geometry oftransmit and receive antenna arrays.

One popular model for MAS channels is the so called clustered channelmodel, that defines groups of scatterers as clusters located around thetransmitters and receivers. In general, the more clusters and the largertheir angular spread, the higher spatial diversity and capacityachievable over wireless links. Clustered channel models have beenvalidated through practical measurements [1-2] and variations of thosemodels have been adopted by different indoor (i.e., IEEE 802.11nTechnical Group [3] for WLAN) and outdoor (3GPP Technical SpecificationGroup for 3G cellular systems [4]) wireless standards.

Other factors that determine the spatial diversity in wireless channelsare the characteristics of the antenna arrays, including: antennaelement spacing [5-7], number of antennas [8-9], array aperture [10-11],array geometry [5,12,13], polarization and antenna pattern [14-28].

A unified model describing the effects of antenna array design as wellas the characteristics of the propagation channel on the spatialdiversity (or degrees of freedom) of wireless links was presented in[29]. The received signal model in [29] is given byy(q)=∫C(q,p)x(p)dp+z(q)where x(p)∈C³ is the polarized vector describing the transmit signal, p,q∈R³ are the polarized vector positions describing the transmit andreceive arrays, respectively, and C(⋅,⋅)∈C^(3×3) is the matrixdescribing the system response between transmit and receive vectorpositions given byC(q,p)=∫∫A _(r)(q,{circumflex over (m)})H({circumflex over(m)},{circumflex over (n)})A _(t)({circumflex over (n)},p)d{circumflexover (n)}d{circumflex over (m)}where A_(t)(⋅,⋅), A_(r)(⋅,⋅)∈C^(3×3) are the transmit and receive arrayresponses respectively and H({circumflex over (m)},{circumflex over(n)})∈C^(3×3) is the channel response matrix with entries being thecomplex gains between transmit direction {circumflex over (n)} andreceive direction {circumflex over (m)}. In DIDO systems, user devicesmay have single or multiple antennas. For the sake of simplicity, weassume single antenna receivers with ideal isotropic patterns andrewrite the system response matrix asC(q,p)=∫H(q,{circumflex over (n)})A({circumflex over (n)},p)d{circumflexover (n)}where only the transmit antenna pattern A({circumflex over (n)}, p) isconsidered.

From the Maxwell equations and the far-field term of the Green function,the array response can be approximated as [29]

${A( {\hat{n},p} )} = {\frac{j\;\eta\; e^{j\; 2\pi\; d_{o}}}{2\lambda^{2}d_{o}}( {I - {\hat{n}\;{\hat{n}}^{H}}} ){a( {\hat{n},p} )}}$with p∈P, P is the space that defines the antenna array and wherea({circumflex over (n)},p)=exp(−j2π{circumflex over (n)} ^(H) p)with ({circumflex over (n)}, p)∈Ω×P. For unpolarized antennas, studyingthe array response is equivalent to study the integral kernel above.Hereafter, we show closed for expressions of the integral kernels fordifferent types of arrays.Unpolarized Linear Arrays

For unpolarized linear arrays of length L (normalized by the wavelength)and antenna elements oriented along the z-axis and centered at theorigin, the integral kernel is given by [29]a(cos θ,p _(z))=exp(−j2πp _(z) cos θ).

Expanding the above equation into a series of shifted dyads, we obtainthat the sinc function have resolution of 1/L and the dimension of thearray-limited and approximately wavevector-limited subspace (i.e.,degrees of freedom) isD _(F) =L|Ω _(θ)|where Ω_(θ)={cos θ: θ∈Θ}. We observe that for broadside arrays|Ω_(θ)|=|Θ| whereas for endfire |Ω_(θ)|≈|Θ|²/2.Unpolarized Spherical Arrays

The integral kernel for a spherical array of radius R (normalized by thewavelength) is given by [29]a({circumflex over (n)},p)=exp{−j2πR[sin θ sin θ′ cos(ϕ−ϕ′)+cos θ cosθ′]}.

Decomposing the above function with sum of spherical Bessel functions ofthe first kind we obtain the resolution of spherical arrays is 1/(πR²)and the degrees of freedom are given byD _(F) =A|Ω|=πR ²|Ω|

where A is the area of the spherical array and |Ω|⊂[0,π)×[0,2π).

Areas of Coherence in Wireless Channels

The relation between the resolution of spherical arrays and their area Ais depicted in FIG. 43. The sphere in the middle is the spherical arrayof area A. The projection of the channel clusters on the unit spheredefines different scattering regions of size proportional to the angularspread of the clusters. The area of size 1/A within each cluster, whichwe call “area of coherence”, denotes the projection of the basisfunctions of the radiated field of the array and defines the resolutionof the array in the wavevector domain.

Comparing FIG. 43 with FIG. 44, we observe that the size of the area ofcoherence decreases as the inverse of the size of the array. In fact,larger arrays can focus energy into smaller areas, yielding largernumber of degrees of freedom DF. Note that to total number of degrees offreedom depends also on the angular spread of the cluster, as shown inthe definition above.

FIG. 45 depicts another example where the array size covers even largerarea than FIG. 44, yielding additional degrees of freedom. In DIDOsystems, the array aperture can be approximated by the total areacovered by all DIDO transmitters (assuming antennas are spaced fractionsof wavelength apart). Then FIG. 45 shows that DIDO systems can achieveincreasing numbers of degrees of freedom by distributing antennas inspace, thereby reducing the size of the areas of coherence. Note thatthese figures are generated assuming ideal spherical arrays. Inpractical scenarios, DIDO antennas spread random across wide areas andthe resulting shape of the areas of coherence may not be as regular asin the figures.

FIG. 46 shows that, as the array size increases, more clusters areincluded within the wireless channel as radio waves are scatterered byincreasing number of objects between DIDO transmitters. Hence, it ispossible to excite an increasing number of basis functions (that spanthe radiated field), yielding additional degrees of freedom, inagreement with the definition above.

The multi-user (MU) multiple antenna systems (MAS) described in thispatent application exploit the area of coherence of wireless channels tocreate multiple simultaneous independent non-interfering data streams todifferent users. For given channel conditions and user distribution, thebasis functions of the radiated field are selected to create independentand simultaneous wireless links to different users in such a way thatevery user experiences interference-free links. As the MU-MAS is awareof the channel between every transmitter and every user, the precodingtransmission is adjusted based on that information to create separateareas of coherence to different users.

In one embodiment of the invention, the MU-MAS employs non-linearprecoding, such as dirty-paper coding (DPC) [30-31] orTomlinson-Harashima (TH) [32-33] precoding. In another embodiment of theinvention, the MU-MAS employs non-linear precoding, such as blockdiagonalization (BD) as in our previous patent applications [0003-0009]or zero-forcing beamforming (ZF-BF) [34].

To enable precoding, the MU-MAS requires knowledge of the channel stateinformation (CSI). The CSI is made available to the MU-MAS via afeedback channel or estimated over the uplink channel, assuminguplink/downlink channel reciprocity is possible in time division duplex(TDD) systems. One way to reduce the amount of feedback required forCSI, is to use limited feedback techniques [35-37]. In one embodiment,the MU-MAS uses limited feedback techniques to reduce the CSI overheadof the control channel. Codebook design is critical in limited feedbacktechniques. One embodiment defines the codebook from the basis functionsthat span the radiated field of the transmit array.

As the users move in space or the propagation environment changes overtime due to mobile objects (such as people or cars), the areas ofcoherence change their locations and shape. This is due to well knowDoppler effect in wireless communications. The MU-MAS described in thispatent application adjusts the precoding to adapt the areas of coherenceconstantly for every user as the environment changes due to Dopplereffects. This adaptation of the areas of coherence is such to createsimultaneous non-interfering channels to different users.

Another embodiment of the invention adaptively selects a subset ofantennas of the MU-MAS system to create areas of coherence of differentsizes. For example, if the users are sparsely distributed in space(i.e., rural area or times of the day with low usage of wirelessresources), only a small subset of antennas is selected and the size ofthe area of coherence are large relative to the array size as in FIG.43. Alternatively, in densely populated areas (i.e., urban areas or timeof the day with peak usage of wireless services) more antennas areselected to create small areas of coherence for users in direct vicinityof each other.

In one embodiment of the invention, the MU-MAS is a DIDO system asdescribed in previous patent applications [0003-0009]. The DIDO systemuses linear or non-linear precoding and/or limited feedback techniquesto create area of coherence to different users.

Numerical Results

We begin by computing the number of degrees of freedom in conventionalmultiple-input multiple-output (MIMO) systems as a function of the arraysize. We consider unpolarized linear arrays and two types of channelmodels: indoor as in the IEEE 802.11n standard for WiFi systems andoutdoor as in the 3GPP-LTE standard for cellular systems. The indoorchannel mode in [3] defines the number of clusters in the range [2, 6]and angular spread in the range [15°, 40° ]. The outdoor channel modelfor urban micro defines about 6 clusters and the angular spread at thebase station of about 20°.

FIG. 47 shows the degrees of freedom of MIMO systems in practical indoorand outdoor propagation scenarios. For example, considering lineararrays with ten antennas spaced one wavelength apart, the maximumdegrees of freedom (or number of spatial channels) available over thewireless link is limited to about 3 for outdoor scenarios and 7 forindoor. Of course, indoor channels provide more degrees of freedom dueto the larger angular spread.

Next we compute the degrees of freedom in DIDO systems. We consider thecase where the antennas distributed over 3D space, such as downtownurban scenarios where DIDO access points may be distributed on differentfloors of adjacent building. As such, we model the DIDO transmitantennas (all connected to each other via fiber or DSL backbone) as aspherical array. Also, we assume the clusters are uniformly distributedacross the solid angle.

FIG. 48 shows the degrees of freedom in DIDO systems as a function ofthe array diameter. We observe that for a diameter equal to tenwavelengths, about 1000 degrees of freedom are available in the DIDOsystem. In theory, it is possible to create up to 1000 non-interferingchannels to the users. The increased spatial diversity due todistributed antennas in space is the key to the multiplexing gainprovided by DIDO over conventional MIMO systems.

As a comparison, we show the degrees of freedom achievable in suburbanenvironments with DIDO systems. We assume the clusters are distributedwithin the elevation angles [α, π−α], and define the solid angle for theclusters as |Ω|=4π cos α. For example, in suburban scenarios withtwo-story buildings, the elevation angle of the scatterers can be α=60°.In that case, the number of degrees of freedom as a function of thewavelength is shown in FIG. 48.

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The growing demand for high-speed wireless services and the increasingnumber of cellular telephone subscribers has produced a radicaltechnology revolution in the wireless industry over the past threedecades from initial analog voice services (AMPS [1-2]) to standardsthat support digital voice (GSM [3-4], IS-95 CDMA [5]), data traffic(EDGE [6], EV-DO [7]) and Internet browsing (WiFi [8-9], WiMAX [10-11],3G [12-13], 4G [14-15]). This wireless technology growth throughout theyears has been enabled by two major efforts:

-   i) The federal communications commission (FCC) [16] has been    allocating new spectrum to support new emerging standards. For    example, in the first generation AMPS systems the number of channels    allocated by the FCC grew from the initial 333 in 1983 to 416 in the    late 1980s to support the increasing number of cellular clients.    More recently, the commercialization of technologies like Wi-Fi,    Bluetooth and ZigBee has been possible with the use of the    unlicensed ISM band allocated by the FCC back in 1985 [17].-   ii) The wireless industry has been producing new technologies that    utilize the limited available spectrum more efficiently to support    higher data rate links and increased numbers of subscribers. One big    revolution in the wireless world was the migration from the analog    AMPS systems to digital D-AMPS and GSM in the 1990s, that enabled    much higher call volume for a given frequency band due to improved    spectral efficiency. Another radical shift was produced in the early    2000s by spatial processing techniques such as multiple-input    multiple-output (MIMO), yielding 4× improvement in data rate over    previous wireless networks and adopted by different standards (i.e.,    IEEE 802.11n for Wi-Fi, IEEE 802.16 for WiMAX, 3GPP for 4G-LTE).

Despite efforts to provide solutions for high-speed wirelessconnectivity, the wireless industry is facing new challenges: to offerhigh-definition (HD) video streaming to satisfy the growing demand forservices like gaming and to provide wireless coverage everywhere(including rural areas, where building the wireline backbone is costlyand impractical). Currently, the most advanced wireless standard systems(i.e., 4G-LTE) cannot provide data rate requirements and latencyconstraints to support HD streaming services, particularly when thenetwork is overloaded with a high volume of concurrent links. Onceagain, the main drawbacks have been the limited spectrum availabilityand lack of spectrally efficient technologies that can truly enhancedata rate and provide complete coverage.

A new technology has emerged in recent years called distributed-inputdistributed-output (DIDO) [18-21] and described in our previous patentapplications [0002-0009]. DIDO technology promises orders of magnitudeincrease in spectral efficiency, making HD wireless streaming servicespossible in overloaded networks.

At the same time, the US government has been addressing the issue ofspectrum scarcity by launching a plan that will free 500 MHz of spectrumover the next 10 years. This plan was released on Jun. 28, 2010 with thegoal of allowing new emerging wireless technologies to operate in thenew frequency bands and providing high-speed wireless coverage in urbanand rural areas [22]. As part of this plan, on Sep. 23, 2010 the FCCopened up about 200 MHz of the VHF and UHF spectrum for unlicensed usecalled “white spaces” [23]. One restriction to operate in thosefrequency bands is that harmful interference must not be created withexisting wireless microphone devices operating in the same band. Assuch, on Jul. 22, 2011 the IEEE 802.22 working group finalized thestandard for a new wireless system employing cognitive radio technology(or spectrum sensing) with the key feature of dynamically monitoring thespectrum and operating in the available bands, thereby avoiding harmfulinterference with coexisting wireless devices [24]. Only recently hasthere been debates to allocate part of the white spaces to licensed useand open it up to spectrum auction [25].

The coexistence of unlicensed devices within the same frequency bandsand spectrum contention for unlicensed versus licensed use have been twomajor issues for FCC spectrum allocation plans throughout the years. Forexample, in white spaces, coexistence between wireless microphones andwireless communications devices has been enabled via cognitive radiotechnology. Cognitive radio, however, can provide only a fraction of thespectral efficiency of other technologies using spatial processing likeDIDO. Similarly, the performance of Wi-Fi systems have been degradingsignificantly over the past decade due to increasing number of accesspoints and the use of Bluetooth/ZigBee devices that operate in the sameunlicensed ISM band and generate uncontrolled interference. Oneshortcoming of the unlicensed spectrum is unregulated use of RF devicesthat will continue to pollute the spectrum for years to come. RFpollution also prevents the unlicensed spectrum from being used forfuture licensed operations, thereby limiting important marketopportunities for wireless broadband commercial services and spectrumauctions.

We propose a new system and methods that allow dynamic allocation of thewireless spectrum to enable coexistence and evolution of differentservices and standards. One embodiment of our method dynamically assignsentitlements to RF transceivers to operate in certain parts of thespectrum and enables obsolescence of the same RF devices to provide:

-   i) Spectrum reconfigurability to enable new types of wireless    operations (i.e., licensed vs. unlicensed) and/or meet new RF power    emission limits. This feature allows spectrum auctions whenever is    necessary, without need to plan in advance for use of licensed    versus unlicensed spectrum. It also allows transmit power levels to    be adjusted to meet new power emission levels enforced by the FCC.-   ii) Coexistence of different technologies operating in the same band    (i.e., white spaces and wireless microphones, WiFi and    Bluetooth/ZigBee) such that the band can be dynamically reallocated    as new technologies are created, while avoiding interference with    existing technologies.-   iii) Seamless evolution of wireless infrastructure as systems    migrate to more advanced technologies that can offer higher spectral    efficiency, better coverage and improved performance to support new    types of services demanding higher QoS (i.e., HD video streaming).

Hereafter, we describe a system and method for planned evolution andobsolescence of a multiuser spectrum. One embodiment of the systemconsists of one or multiple centralized processors (CP) 4901-4904 andone or multiple distributed nodes (DN) 4911-4913 that communicate viawireline or wireless connections as depicted in FIG. 49. For example, inthe context of 4G-LTE networks [26], the centralized processor is theaccess core gateway (ACGW) connected to several Node B transceivers. Inthe context of Wi-Fi, the centralized processor is the internet serviceprovider (ISP) and the distributed nodes are Wi-Fi access pointsconnected to the ISP via modems or direct connection to cable or DSL. Inanother embodiment of the invention, the system is a distributed-inputdistributed-output (DIDO) system [0002-0009] with one centralizedprocessor (or BTS) and distributed nodes being the DIDO access points(or DIDO distributed antennas connected to the BTS via the BSN).

The DNs 4911-4913 communicate with the CPs 4901-4904. The informationexchanged from the DNs to the CP is used to dynamically adjust theconfiguration of the nodes to the evolving design of the networkarchitecture. In one embodiment, the DNs 4911-4913 share theiridentification number with the CP. The CP store the identificationnumbers of all DNs connected through the network into lookup tables orshared database. Those lookup tables or database can be shared withother CPs and that information is synchronized such that all CPs havealways access to the most up to date information about all DNs on thenetwork.

For example, the FCC may decide to allocate a certain portion of thespectrum to unlicensed use and the proposed system may be designed tooperate within that spectrum. Due to scarcity of spectrum, the FCC maysubsequently need to allocate part of that spectrum to licensed use forcommercial carriers (i.e., AT&T, Verizon, or Sprint), defense, or publicsafety. In conventional wireless systems, this coexistence would not bepossible, since existing wireless devices operating in the unlicensedband would create harmful interference to the licensed RF transceivers.In our proposed system, the distributed nodes exchange controlinformation with the CPs 4901-4903 to adapt their RF transmission to theevolving band plan. In one embodiment, the DNs 4911-4913 were originallydesigned to operate over different frequency bands within the availablespectrum. As the FCC allocates one or multiple portions of that spectrumto licensed operation, the CPs exchange control information with theunlicensed DNs and reconfigure them to shut down the frequency bands forlicensed use, such that the unlicensed DNs do not interfere with thelicensed DNs. This scenario is depicted in FIG. 50 where the unlicensednodes (e.g., 5002) are indicated with solid circles and the licensednodes with empty circles (e.g., 5001). In another embodiment, the wholespectrum can be allocated to the new licensed service and the controlinformation is used by the CPs to shut down all unlicensed DNs to avoidinterference with the licensed DNs. This scenario is shown in FIG. 51where the obsolete unlicensed nodes are covered with a cross.

By way of another example, it may be necessary to restrict poweremissions for certain devices operating at given frequency band to meetthe FCC exposure limits [27]. For instance, the wireless system mayoriginally be designed for fixed wireless links with the DNs 4911-4913connected to outdoor rooftop transceiver antennas. Subsequently, thesame system may be updated to support DNs with indoor portable antennasto offer better indoor coverage. The FCC exposure limits of portabledevices are more restrictive than rooftop transmitters, due to possiblycloser proximity to the human body. In this case, the old DNs designedfor outdoor applications can be reused for indoor applications as longas the transmit power setting is adjusted. In one embodiment of theinvention the DNs are designed with predefined sets of transmit powerlevels and the CPs 4901-4903 send control information to the DNs4911-4913 to select new power levels as the system is upgraded, therebymeeting the FCC exposure limits. In another embodiment, the DNs aremanufactured with only one power emission setting and those DNsexceeding the new power emission levels are shut down remotely by theCP.

In one embodiment, the CPs 4901-4903 monitor periodically all DNs4911-4913 in the network to define their entitlement to operate as RFtransceivers according to a certain standard. Those DNs that are not upto date can be marked as obsolete and removed from the network. Forexample, the DNs that operate within the current power limit andfrequency band are kept active in the network, and all the others areshut down. Note that the DN parameters controlled by the CP are notlimited to power emission and frequency band; it can be any parameterthat defines the wireless link between the DN and the client devices.

In another embodiment of the invention, the DNs 4911-4913 can bereconfigured to enable the coexistence of different standard systemswithin the same spectrum. For example, the power emission, frequencyband or other configuration parameters of certain DNs operating in thecontext of WLAN can be adjusted to accommodate the adoption of new DNsdesigned for WPAN applications, while avoiding harmful interference.

As new wireless standards are developed to enhance data rate andcoverage in the wireless network, the DNs 4911-4913 can be updated tosupport those standards. In one embodiment, the DNs are software definedradios (SDR) equipped with programmable computational capability such asFPGA, DSP, CPU, GPU and/or GPGPU that run algorithms for baseband signalprocessing. If the standard is upgraded, new baseband algorithms can beremotely uploaded from the CP to the DNs to reflect the new standard.For example, in one embodiment the first standard is CDMA-based andsubsequently it is replaced by OFDM technology to support differenttypes of systems. Similarly, the sample rate, power and other parameterscan be updated remotely to the DNs. This SDR feature of the DNs allowsfor continuous upgrades of the network as new technologies are developedto improve overall system performance.

In another embodiment, the system described herein is a cloud wirelesssystem consisting of multiple CPs, distributed nodes and a networkinterconnecting the CPs to the DNs. FIG. 52 shows one example of cloudwireless system where the nodes identified with solid circles (e.g.,5203) communicate to CP 5206, the nodes identified with empty circlescommunicate to CP 5205 and the CPs 5205-5206 communicate between eachother all through the network 5201. In one embodiment of the invention,the cloud wireless system is a DIDO system and the DNs are connected tothe CP and exchange information to reconfigure periodically or instantlysystem parameters, and dynamically adjust to the changing conditions ofthe wireless architecture. In the DIDO system, the CP is the DIDO BTS,the distributed nodes are the DIDO distributed antennas, the network isthe BSN and multiple BTSs are interconnected with each other via theDIDO centralized processor as described in our previous patentapplications [0002-0009].

All DNs 5202-5203 within the cloud wireless system can be grouped indifferent sets. These sets of DNs can simultaneously createnon-interfering wireless links to the multitude of client devices, whileeach set supporting a different multiple access techniques (e.g., TDMA,FDMA, CDMA, OFDMA and/or SDMA), different modulations (e.g., QAM, OFDM)and/or coding schemes (e.g., convolutional coding, LDPC, turbo codes).Similarly, every client may be served with different multiple accesstechniques and/or different modulation/coding schemes. Based on theactive clients in the system and the standard they adopt for theirwireless links, the CPs 5205-5206 dynamically select the subset of DNsthat can support those standards and that are within range of the clientdevices.

REFERENCE

-   [1] Wikipedia, “Advanced Mobile Phone System”    http://en.wikipedia.org/wiki/Advanced_Mobile_Phone_System-   [2] AT&T, “1946: First Mobile Telephone Call”    http://www.corp.att.com/attlabs/reputation/timeline/46mobile.html-   [3] GSMA, “GSM technology”    http://www.gsmworld.com/technology/index.htm-   [4] ETSI, “Mobile technologies GSM”    http://www.etsi.org/WebSite/Technologies/gsm.aspx-   [5] Wikipedia, “IS-95” http://en.wikipedia.org/wiki/IS-95-   [6] Ericsson, “The evolution of EDGE”    http://www.ericsson.com/res/docs/whitepapers/evolution_to_edge.pdf-   [7] Q. Bi (2004-03). “A Forward Link Performance Study of the    1×EV-DO Rel. 0 System Using Field Measurements and Simulations”    (PDF). Lucent Technologies.    http://www.cdg.org/resources/white_papers/files/Lucent %201×EV-DO    %20Rev %20O %20Mar %2004.pdf-   [8] Wi-Fi alliance, http://www.wi-fi.org/-   [9] Wi-Fi alliance, “Wi-Fi certified makes it Wi-Fi”    http://www.wi-fi.org/files/WFA_Certification_Overview_WP_en.pdf-   [10] WiMAX forum, http://www.wimaxforum.org/-   [11] C. Eklund, R. B. Marks, K. L. Stanwood and S. Wang, “IEEE    Standard 802.16: A Technical Overview of the WirelessMAN™ Air    Interface for Broadband Wireless Access”    http://ieee802.org/16/docs/02/C80216-02_05. pdf-   [12] 3GPP, “UMTS”, http://www.3gpp.org/article/umts-   [13] H. Ekstrom, A. Furuskär, J. Karlsson, M. Meyer, S. Parkvall, J.    Torsner, and M. Wahlqvist “Technical Solutions for the 3G Long-Term    Evolution”, IEEE Communications Magazine, pp. 38-45, March 2006-   [14] 3GPP, “LTE”, http://www.3gpp.org/LTE-   [15] Motorola, “Long Term Evolution (LTE): A Technical Overview”,    http://business.motorola.com/experiencelte/pdf/LTETechnicalOverview.pdf-   [16] Federal Communications Commission, “Authorization of Spread    Spectrum Systems Under Parts 15 and 90 of the FCC Rules and    Regulations”, June 1985.-   [17] ITU, “ISM band”    http://www.itu.int/ITU-R/terrestrial/faq/index.html#g013-   [18] S. Perlman and A. Forenza “Distributed-input distributed-output    (DIDO) wireless technology: a new approach to multiuser wireless”,    August 2011 http://www.rearden.com/DIDO/DIDO_White_Paper_110727.pdf-   [19] Bloomberg Businessweek, “Steve Perlman's Wireless Fix”, Jul.    27, 2011    http://www.businessweek.com/magazine/the-edison-of-silicon-valley-07272011.html-   [20] Wired, “Has OnLive's Steve Perlman Discovered Holy Grail of    Wireless?”, Jun. 30, 2011    http://www.wired.com/epicenter/2011/06/perlman-holy-grail-wireless/-   [21] The Wall Street Journal “Silicon Valley Inventor's Radical    Rewrite of Wireless”, Jul. 28, 2011    http://blogs.wsj.com/digits/2011/07/28/silicon-valley-inventors-radical-rewrite-of-wireless/-   [22] The White House, “Presidential Memorandum: Unleashing the    Wireless Broadband Revolution”, Jun. 28, 2010    http://www.whitehouse.gov/the-press-office/presidential-memorandum-unleashing-wireless-broadband-revolution-   [23] FCC, “Open commission meeting”, Sep. 23, 2010    http://reboot.fcc.gov/open-meetings/2010/september-   [24] IEEE 802.22, “IEEE 802.22 Working Group on Wireless Regional    Area Networks”, http://www.ieee802.org/22/-   [25] “A bill”, 112th congress, 1^(st) session, Jul. 12, 2011    http://republicans.energycommerce.house.gov/Media/file/Hearings/Telecom/071511/DiscussionDraft.pdf-   [26] H. Ekström, A. Furuskär, J. Karlsson, M. Meyer, S. Parkvall, J.    Torsner, and M. Wahlqvist “Technical Solutions for the 3G Long-Term    Evolution”, IEEE Communications Magazine, pp. 38-45, March 2006-   [27] FCC, “Evaluating compliance with FCC guidelines for human    exposure to radiofrequency electromagnetic fields,” OET Bulletin 65,    Edition 97-01, August 1997    V. System and Method to Compensate for Doppler Effects in    Distributed-Input Distributed-Output Wireless Systems

In this portion of the detailed description we describe a multiuser (MU)multiple antenna system (MAS) for multiuser wireless transmissions thatadaptively reconfigures its parameters to compensate for Doppler effectsdue to user mobility or changes in the propagation environment. In oneembodiment, the MAS is a distributed-input distributed-output (DIDO)system as described the co-pending patent applications [0002-0016] anddepicted in FIG. 53. The DIDO system of one embodiment includes thefollowing components:

-   -   User Equipment (UE): The UE 5301 of one embodiment includes an        RF transceiver for fixed or mobile clients receiving data        streams over the downlink (DL) channel from the DIDO backhaul        and transmitting data to the DIDO backhaul via the uplink (UL)        channel    -   Base Transceiver Station (BTS): The BTSs 5310-5314 of one        embodiment interface the DIDO backhaul with the wireless        channel. BTSs 5310-5314 are access points consisting of DAC/ADC        and radio frequency (RF) chain to convert the baseband signal to        RF. In some cases, the BTS is a simple RF transceiver equipped        with power amplifier/antenna and the RF signal is carried to the        BTS via RF-over-fiber technology as described in our patent        application [0010].    -   Controller (CTR): The CTR 5320 in one embodiment is one        particular type of BTS designed for certain specialized features        such as transmitting training signals for time/frequency        synchronization of the BTSs and/or the UEs,        receiving/transmitting control information from/to the UEs,        receiving the channel state information (CSI) or channel quality        information from the UEs.    -   Centralized Processor (CP): The CP 5340 of one embodiment is a        DIDO server interfacing the Internet or other types of external        networks 5350 with the DIDO backhaul. The CP computes the DIDO        baseband processing and sends the waveforms to the distributed        BTSs for DL transmission    -   Base Station Network (BSN): The BSN 5330 of one embodiment is        the network connecting the CP to the distributed BTSs carrying        information for either the DL or the UL channel. The BSN is a        wireline or a wireless network or a combination of the two. For        example, the BSN is a DSL, cable, optical fiber network, or        line-of-sight or non-line-of-sight wireless link. Furthermore,        the BSN is a proprietary network, or a local area network, or        the Internet.

As described in the co-pending applications, the DIDO system createsindependent channels to multiple users, such that each user receivesinterference-free channels. In DIDO systems, this is achieved byemploying distributed antennas or BTSs to exploit spatial diversity. Inone embodiment, the DIDO system exploits spatial, polarization and/orpattern diversity to increase the degrees of freedom within eachchannel. The increased degrees of freedom of the wireless link are usedto transmit independent data streams to an increased number of UEs(i.e., multiplexing gain) and/or improve coverage (i.e., diversitygain).

The BTSs 5310-5314 are placed anywhere that is convenient where there isaccess to the Internet or BSN. In one embodiment of the invention, theUEs 5301-5305 are placed randomly between, around and/or surrounded bythe BTSs or distributed antennas as depicted in FIG. 54.

In one embodiment, the BTSs 5310-5314 send a training signal and/orindependent data streams to the UEs 5301 over the DL channel as depictedin FIG. 55. The training signal is used by the UEs for differentpurposes, such as time/frequency synchronization, channel estimationand/or estimation of the channel state information (CSI). In oneembodiment of the invention, the MU-MAS DL employs non-linear precoding,such as dirty-paper coding (DPC) [1-2] or Tomlinson-Harashima (TH) [3-4]precoding. In another embodiment of the invention, the MU-MAS DL employsnon-linear precoding, such as block diagonalization (BD) as described inthe co-pending patent applications [0003-0009] or zero-forcingbeamforming (ZF-BF) [5]. If the number of BTSs is larger than the UEs,the extra BTSs are used to increase link quality to every UE viadiversity schemes such as antenna selection or eigenmode selectiondescribed in [0002-0016]. If the number of BTSs is smaller than the UEs,the extra UEs share the wireless links with the other UEs viaconventional multiplexing techniques (e.g., TDMA, FDMA, CDMA, OFDMA).

The UL channel is used to transmit data from the UEs 5301 to the CP 5340and/or the CSI (or channel quality information) employed by the DIDOprecoder. In one embodiment, the UL channels from the UEs aremultiplexed via conventional multiplexing techniques (e.g., TDMA, FDMA,CDMA, OFDMA) to the CTR as depicted in FIG. 56 or to the closest BTS. Inanother embodiment of the invention, spatial processing techniques areused to separate the UL channels from the UEs 5301 to the distributedBTSs 5310-5314 as depicted in FIG. 57. For example, UL streams aretransmitted from the client to the DIDO antennas via multiple-inputmultiple-output (MIMO) multiplexing schemes. The MIMO multiplexingschemes include transmitting independent data streams from the clientsand using linear or non-linear receivers at the DIDO antennas to removeco-channel interference. In another embodiment, the downlink weights areused over the uplink to demodulate the uplink streams, assuming UL/DLchannel reciprocity holds and the channel does not vary significantlybetween DL and UL transmission due to Doppler effects. In anotherembodiment, a maximum ratio combining (MRC) receiver is used over the ULchannel to increase signal quality at the DIDO antennas from everyclient.

The data, control information and CSI sent over the DL/UL channels isshared between the CP 5340 and the BTSs 5310-5314 via the BSN 5330. Theknown training signals for the DL channel can be stored in memory at theBTSs 5310-5314 to reduce overhead over the BSN 5330. Depending on thetype of network (i.e., wireless versus wireline, DSL versus cable orfiber optic), there may not be a sufficient data rate available over theBSN 5330 to exchange information between the CP 5340 and the BTSs5310-5314, especially when the baseband signal is delivered to the BTSs.For example, let us assume the BTSs transmit 10 Mbps independent datastreams to every UE over 5 MHz bandwidth (depending on the digitalmodulation and FEC coding scheme used over the wireless link). If 16bits of quantization are used for the real and 16 for the imaginarycomponents, the baseband signal requires 160 Mbps of data throughputfrom the CP to the BTSs over the BSN. In one embodiment, the CP and theBTSs are equipped with encoders and decoders to compress and decompressinformation sent over the BSN. In the forward link, the precodedbaseband data sent from the CP to the BTSs is compressed to reduce theamount of bits and overhead sent over the BSN. Similarly, in the reverselink, the CSI as well as data (sent over the uplink channel from the UEsto the BTSs) are compressed before being transmitted over the BSN fromthe BTSs to the CP. Different compression algorithms are employed toreduce the amount of bits and overhead sent over the BSN, including butnot limited to lossless and/or lossy techniques [6].

One feature of DIDO systems employed in one embodiment is making the CP5340 aware of the CSI or channel quality information between all BTSs53105314 and UEs 5301 to enable precoding. As explained in [0006], theperformance of DIDO depends on the rate at which the CSI is delivered tothe CP relative to the rate of change of the wireless links. It is wellknown that variations of the channel complex gain are due to UE mobilityand/or changes in the propagation environment that cause Dopplereffects. The rate of change of the channel is measured in terms ofchannel coherence time (T_(c)) that is inversely proportional to themaximum Doppler shift. For DIDO transmissions to perform reliably, thelatency due to CSI feedback must be a fraction (e.g., 1/10 or less) ofthe channel coherence time. In one embodiment, the latency over the CSIfeedback loop is measured as the time between the time at which the CSItraining is sent and the time the precoded data is demodulated at the UEside, as depicted in FIG. 58.

In frequency division duplex (FDD) DIDO systems the BTSs 5310-5314 sendCSI training to the UEs 5301, that estimate the CSI and feedback to theBTSs. Then the BTSs send the CSI via the BSN to the CP 5340, thatcomputes the DIDO precoded data streams and sends those back to the BTSsvia the BSN 5330. Finally the BTSs send precoded streams to the UEs thatdemodulate the data. Referring to FIG. 58, the overall latency for theDIDO feedback loop is given by2*T _(DL) +T _(UL) +T _(BSN) +T _(CP)where T_(DL) and T_(UL) include the times to build, send and process thedownlink and uplink frames, respectively, T_(BSN) is the round-tripdelay over the BSN and T_(CP) is the time taken by the CP to process theCSI, generate the precoded data streams for the UEs and scheduledifferent UEs for the current transmission. In this case, T_(DL) ismultiplied by 2 to account for the training signal time (from the BTS tothe UE) and the feedback signal time (from the UE to the BTS). In timedivision duplex (TDD), if channel reciprocity can be exploited, thefirst step is skipped (i.e., transitting a CSI training signal from theBTS to the UE) as the UEs send CSI training to the BTSs that compute theCSI and send it to the CP. Hence, in this embodiment, the overalllatency for the DIDO feedback loop isT _(DL) +T _(UL) +T _(BSN) +T _(CP)

The latency T_(BSN) depends on the type of BSN whether dedicated cable,DSL, fiber optic connection or general Internet. Typical values may varybetween fractions of 1 msec to 50 msec. The computational time at the CPcan be reduced if the DIDO processing is implemented at the CP ondedicated processors such as ASIC, FPGA, DSP, CPU, GPU and/or GPGPU.Moreover, if the number of BTSs 5310-5314 exceeds the number of UEs5301, all the UEs can be served at the same time, thereby removinglatency due to multiuser scheduling. Hence, the latency T_(CP) isnegligible compared to T_(BSN). Finally, transmit and receive processingfor the DL and UL is typically implemented on ASIC, FPGA or DSP withnegligible computational time and if the signal bandwidth is relativelylarge (e.g. more than 1 MHz) the frame duration can be made very small(i.e., less than 1 msec). Therefore, also T_(DL) and T_(UL) arenegligible compared to T_(BSN).

In one embodiment of the invention, the CP 5340 tracks the Dopplervelocity of all UEs 5301 and dynamically assigns the BTSs 5310-5314 withthe lowest T_(BSN) to the UEs with higher Doppler. This adaptation isbased on different criteria:

-   -   Type of BSN: For example, dedicated fiber optic links typically        experience lower latency than cable modems or DSL. Then the        lower latency BSNs are used for high-mobility UEs (e.g., cars on        freeways, trains), whereas the higher-latency BSNs are used for        the fixed-wireless or low-mobility UEs (e.g., home equipment,        pedestrians, cars in residential areas)    -   Type of QoS: For example, the BSN can support different types of        DIDO or non-DIDO traffic. It is possible to define quality of        service (QoS) with different priorities for different types of        traffic. For example, the BSN assigns high priority to DIDO        traffic and low priority to non-DIDO traffic. Alternatively,        high priority QoS is assigned to traffic for high-mobility UEs        and low priority QoS to UEs with low-mobility.    -   Long-term statistics: For example, the traffic over the BSN may        vary significantly depending on the time of the day (e.g., night        use for homes and day use for offices). Higher traffic load may        result in higher latency. Then, in different times of the day,        the BSNs with higher traffic, if it results in higher latency,        are used for low-mobility UEs, whereas the BSNs with lower        traffic, if it results in lower latency, are used for the        high-mobility UEs    -   Short-term statistics: For example, any BSN can be affected by        temporary network congestion that can result in higher latency.        Then the CP can adaptively select the BTSs from congested BSNs,        if the congestion cause higher latency, for the low-mobility UEs        and the remaining BSNs, if they are lower latency, for the        high-mobility UEs.

In another embodiment of the invention, the BTSs 5310-5314 are selectedbased on the Doppler experienced on each individual BTS-UE link. Forexample, in the line-of-sight (LOS) link B in FIG. 59, the maximumDoppler shift is a function of the angle (ϕ) between the BTS-UE link andthe vehicular velocity (v), according to the well known equation

$f_{d} = {{\frac{v}{\lambda} \cdot \cos}\;\phi}$where λ is the wavelength corresponding to the carrier frequency. Hence,in LOS channels the Doppler shift is maximum for link A and nearly zerofor link C in FIG. 59. In non-LOS (NLOS) the maximum Doppler shiftdepends on the direction of the multipaths around the UEs, but ingeneral because of the distributed nature of the BTSs in DIDO systems,some BTSs will experience higher Doppler for a given UE (e.g., BTS 5312)whereas other BTSs will experience lower Doppler for that given UE(e.g., BTS 5314).

In one embodiment, the CP tracks the Doppler velocity over every BTS-UElink and selects only the links with the lowest Doppler effect for everyUE. Similarly to the techniques described in [0002], the CP 5340 definesthe “user cluster” for every UE 5301. The user cluster is the set ofBTSs with good link quality (defined based on certain signal-to-noiseratio, SNR, threshold) to the UE and low Doppler (defined, for example,based on a predefined Doppler threshold) as depicted in FIG. 60. In FIG.60, BTSs 5 through 10 all have good SNR to the UE1, but only BTSs 6through 9 experience low Doppler effect (e.g., below the specifiedthreshold).

The CP of this embodiment records all of the values of SNR and Dopplerfor every BTS-UE link into a matrix and for each UE it selects thesubmatrix that satisfies the SNR and Doppler thresholds. In the exampledepicted in FIG. 61, the submatrix is identified by the green dottedline surrounding C_(2,6), C_(2,7), C_(3,9), C_(4,7), C_(4,8), C_(4,9),and C_(5,6). DIDO precoding weights are computed for that UE based onthat submatrix. Note that BTSs 5 and 10 are reachable by UEs 2, 3, 4, 5and 7 as shown in the table in FIG. 61. Then, to avoid interference toUE1 when transmitting to those other UEs, the BTSs 5 and 10 either mustswitched off or assigned to different orthogonal channels based onconventional multiplexing techniques such as TDMA, FDMA, CDMA or OFDMA.

In another embodiment, the adverse effect of Doppler on the performanceof DIDO precoding systems is reduced via linear prediction, which is onetechnique to estimate the complex channel coefficients in the futurebased on past channel estimates. By way of example and not limitation,different prediction algorithms for single-input single-output (SISO)and OFDM wireless systems were proposed in [7-11]. Knowing the futurechannel complex coefficients it is possible to reduce the error due tooutdated CSI. For example, FIG. 62 shows the channel gain (or CSI) atdifferent times: i) t_(CTR) is the time at which the CTR in FIG. 58receives the CSI from the UEs in FDD systems (or equivalently the BTSsestimate the CSI from the UL channel exploiting DL/UL reciprocity in TDDsystems); ii) t_(CP) is the time at which the CSI is delivered to the CPvia the BSN; iii) t_(BTS) is the time at which the CSI is used forprecoding over the wireless link. In FIG. 62 we observe that due to thedelay T_(BSN) (also depicted in FIG. 58), the CSI estimated at timet_(CTR) will be outdated (i.e., complex channel gain has changed) by thetime is used for wireless transmission over the DL channel at timet_(BTS). One way to avoid this effect due to Doppler is to run theprediction method at the CP. The CSI estimates available at the CP attime t_(CTR) is delayed of T_(BSN)/2 due to CTR-to-CP latency andcorresponds to the channel gain at time t₀ in FIG. 62. Then, the CP usesall or part of the CSI estimated before time t₀ and stored in memory topredict the future channel coefficient at time t₀+T_(BSN)=t_(CP). If theprediction algorithm has minimal error propagation, the predicted CSI attime t_(CP) reproduces reliably the channel gain in the future. The timedifference between the predicted CSI and the current CSI is calledprediction horizon and in SISO systems typically scales with the channelcoherence time.

In DIDO systems the prediction algorithm is more complex since itestimates the future channel coefficients both in time and spacedomains. Linear prediction algorithms exploiting spatio-temporalcharacteristics of MIMO wireless channels were described in [12-13]. In[13] it was shown that the performance of the prediction algorithms inMIMO systems (measured in terms of mean squared error, or MSE) improvesfor higher channel coherence time (i.e., reduce Doppler effect) andlower channel coherence distance (due to lower spatial correlation).Hence the prediction horizon (expressed in seconds) of spatial-temporalmethods is directly proportional to the channel coherence time andinversely proportional to the channel coherence distance. In DIDOsystems the coherence distance is low due to high spatial selectivityproduce by the distributed antennas.

Described herein is are prediction techniques that exploit temporal andspatial diversity of DIDO systems to predict the vector channel (i.e.,CSI from the BTSs to the UEs) in the future. These embodiments exploitspatial diversity available in wireless channels to obtain negligibleCSI prediction error and an extended prediction horizon over anyexisting SISO and MIMO prediction algorithms. One important feature ofthese techniques is to exploit distributed antennas given that theyreceive uncorrelated complex channel coefficients from the distributedUEs.

In one embodiment of the invention, the spatial and temporal predictoris combined with estimator in the frequency domain to allow CSIprediction over all the available subcarriers in the system, such as inOFDM systems. In another embodiment of the invention, the DIDO precodingweights are predicted (rather than the CSI) based on previous estimatesof the DIDO weights.

REFERENCES

-   [1] M. Costa, “Writing on dirty paper,” IEEE Transactions on    Information Theory, Vol. 29, No. 3, Page(s): 439-441, May 1983.-   [2] U. Erez, S. Shamai (Shitz), and R. Zamir, “Capacity and    lattice-strategies for cancelling known interference,” Proceedings    of International Symposium on Information Theory, Honolulu, Hi.,    November 2000.-   [3] M. Tomlinson, “New automatic equalizer employing modulo    arithmetic,” Electronics Letters, Page(s): 138-139, March 1971.-   [4] H. Miyakawa and H. Harashima, “A method of code conversion for    digital communication channels with intersymbol interference,”    Transactions of the Institute of Electronic-   [5] R. A. Monziano and T. W. Miller, Introduction to Adaptive    Arrays, New York: Wiley, 1980.-   [6] Guy E. Blelloch, “Introduction to Data Compression”, Carnegie    Mellon University Tech. Report September 2010-   [7] A. Duel-Hallen, S. Hu, and H. Hallen, “Long-Range Prediction of    Fading Signals,” IEEE Signal Processing Mag., vol. 17, no. 3, pp.    62-75, May 2000.-   [8] A. Forenza and R. W. Heath, Jr., “Link Adaptation and Channel    Prediction in Wireless OFDM Systems,” in Proc. IEEE Midwest Symp. on    Circuits and Sys., August 2002, pp. 211-214.-   [9] M. Sternad and D. Aronsson, “Channel estimation and prediction    for adaptive OFDM downlinks [vehicular applications],” in Proc. IEEE    Vehicular Technology Conference, vol. 2, October 2003, pp.    1283-1287.-   [10] D. Schafhuber and G. Matz, “MMSE and Adaptive Prediction of    Time-Varying Channels for OFDM Systems,” IEEE Trans. Wireless    Commun., vol. 4, no. 2, pp. 593-602, March 2005.-   [11] I. C. Wong and B. L. Evans, “Joint Channel Estimation and    Prediction for OFDM Systems,” in Proc. IEEE Global    Telecommunications Conference, St. Louis, Mo., December 2005.-   [12] M. Guillaud and D. Slock, “A specular approach to MIMO    frequencyselective channel tracking and prediction,” in Proc. IEEE    Signal Processing Advances in Wireless Communications, July 2004,    pp. 59-63.-   [13] Wong, I. C. Evans, B. L., “Exploiting Spatio-Temporal    Correlations in MIMO Wireless Channel Prediction”, IEEE Globecom    Conf., pp. 1-5, December 2006

Embodiments of the invention may include various steps as set forthabove. The steps may be embodied in machine-executable instructionswhich cause a general-purpose or special-purpose processor to performcertain steps. For example, the various components within the BaseStations/APs and Client Devices described above may be implemented assoftware executed on a general purpose or special purpose processor. Toavoid obscuring the pertinent aspects of the invention, various wellknown personal computer components such as computer memory, hard drive,input devices, etc., have been left out of the figures.

Alternatively, in one embodiment, the various functional modulesillustrated herein and the associated steps may be performed by specifichardware components that contain hardwired logic for performing thesteps, such as an application-specific integrated circuit (“ASIC”) or byany combination of programmed computer components and custom hardwarecomponents.

In one embodiment, certain modules such as the Coding, Modulation andSignal Processing Logic 903 described above may be implemented on aprogrammable digital signal processor (“DSP”) (or group of DSPs) such asa DSP using a Texas Instruments' TMS320× architecture (e.g., aTMS320C6000, TMS320C5000, . . . etc). The DSP in this embodiment may beembedded within an add-on card to a personal computer such as, forexample, a PCI card. Of course, a variety of different DSP architecturesmay be used while still complying with the underlying principles of theinvention.

Elements of the present invention may also be provided as amachine-readable medium for storing the machine-executable instructions.The machine-readable medium may include, but is not limited to, flashmemory, optical disks, CD-ROMs, DVD ROMs, RAMs, EPROMs, EEPROMs,magnetic or optical cards, propagation media or other type ofmachine-readable media suitable for storing electronic instructions. Forexample, the present invention may be downloaded as a computer programwhich may be transferred from a remote computer (e.g., a server) to arequesting computer (e.g., a client) by way of data signals embodied ina carrier wave or other propagation medium via a communication link(e.g., a modem or network connection).

Throughout the foregoing description, for the purposes of explanation,numerous specific details were set forth in order to provide a thoroughunderstanding of the present system and method. It will be apparent,however, to one skilled in the art that the system and method may bepracticed without some of these specific details. Accordingly, the scopeand spirit of the present invention should be judged in terms of theclaims which follow.

Moreover, throughout the foregoing description, numerous publicationswere cited to provide a more thorough understanding of the presentinvention. All of these cited references are incorporated into thepresent application by reference.

The invention claimed is:
 1. A multiuser (MU) multiple antenna system(MAS) comprising: a plurality of users; a plurality of distributedtransceiver stations or antennas cooperatively precoding a plurality ofdata streams to create a plurality of concurrent non-interfering datalinks with the users; one or more centralized units communicativelycoupled to the plurality of distributed transceiver stations or antennasvia a network; the network comprising wireline or wireless links or acombination of both, employed as a backhaul communication channel; theone or more centralized units communicating with the distributedtransceiver stations or antennas over the network to adaptivelyreconfigure communications between the distributed transceiver stationsor antennas and users to compensate for Doppler effects due to usermobility or changes in the propagation environment.
 2. The system as inclaim 1 wherein the precoded data streams are obtained from precodingweights computed from the channel state information (CSI).
 3. A methodimplemented within a multiuser (MU) multiple antenna system (MAS)comprising: a plurality of users; a plurality of distributed transceiverstations or antennas cooperatively precoding a plurality of data streamsto create a plurality of concurrent non-interfering data links with theusers; one or more centralized units communicatively coupled to theplurality of distributed transceiver stations or antennas via a network;the network comprising wireline or wireless links or a combination ofboth, employed as a backhaul communication channel; the methodcomprising the one or more centralized units communicating with thedistributed transceiver stations or antennas over the network toadaptively reconfigure communications between the distributedtransceiver stations or antennas and users to compensate for Dopplereffects due to user mobility or changes in the propagation environment.4. The method as in claim 3 wherein the centralized unit is acentralized processor (CP), the network is a base station network (BSN)and the distributed transceiver stations are distributed basetransceiver stations (BTSs).
 5. The method as in claim 4 wherein the CPand the BTSs are equipped with encoders/decoders to compress/decompressinformation exchanged among them over the BSN.
 6. The method as in claim4 wherein the CP adaptively selects the BTSs to be used for UEs that arestationary, or move at pedestrian speed, or at vehicle speed based onlatency over the BSN.
 7. The method as in claim 6 wherein the adaptationis based on latency over the BSN, or quality of service (QoS), oraverage traffic statistics (e.g., daytime or nighttime use of differentnetworks), or instantaneous traffic statistics (e.g., temporary networkcongestions) over the BSN.
 8. The method as in claim 4 wherein the CPadaptively selects the BTSs to be used for users that are stationary, ormove at pedestrian speed, or at vehicle speed based on the Dopplerspreads of the BTS-user links
 89. 9. The method as in claim 3 whereinthe precoded data streams are obtained from precoding weights computedfrom the channel state information (CSI).